High step‐up high step‐down bidirectional DC/DC converter
2017; Institution of Engineering and Technology; Volume: 10; Issue: 12 Linguagem: Inglês
10.1049/iet-pel.2016.0977
ISSN1755-4543
AutoresEbrahim Babaei, Zahra Saadatizadeh, Carlo Cecati,
Tópico(s)Multilevel Inverters and Converters
ResumoIET Power ElectronicsVolume 10, Issue 12 p. 1556-1571 Research ArticleFree Access High step-up high step-down bidirectional DC/DC converter Ebrahim Babaei, Corresponding Author Ebrahim Babaei e-babaei@tabrizu.ac.ir Faculty of Electrical and Computer Engineering, University of Tabriz, Tabriz, Iran Engineering Faculty, Near East University, 99138 Nicosia, North Cyprus, Mersin 10, TurkeySearch for more papers by this authorZahra Saadatizadeh, Zahra Saadatizadeh Faculty of Electrical and Computer Engineering, University of Tabriz, Tabriz, IranSearch for more papers by this authorCarlo Cecati, Carlo Cecati Department of Information Engineering, Computer Science and Mathematics, University of L'Aquila, L'Aquila, ItalySearch for more papers by this author Ebrahim Babaei, Corresponding Author Ebrahim Babaei e-babaei@tabrizu.ac.ir Faculty of Electrical and Computer Engineering, University of Tabriz, Tabriz, Iran Engineering Faculty, Near East University, 99138 Nicosia, North Cyprus, Mersin 10, TurkeySearch for more papers by this authorZahra Saadatizadeh, Zahra Saadatizadeh Faculty of Electrical and Computer Engineering, University of Tabriz, Tabriz, IranSearch for more papers by this authorCarlo Cecati, Carlo Cecati Department of Information Engineering, Computer Science and Mathematics, University of L'Aquila, L'Aquila, ItalySearch for more papers by this author First published: 18 August 2017 https://doi.org/10.1049/iet-pel.2016.0977Citations: 36AboutSectionsPDF ToolsRequest permissionExport citationAdd to favoritesTrack citation ShareShare Give accessShare full text accessShare full-text accessPlease review our Terms and Conditions of Use and check box below to share full-text version of article.I have read and accept the Wiley Online Library Terms and Conditions of UseShareable LinkUse the link below to share a full-text version of this article with your friends and colleagues. Learn more.Copy URL Share a linkShare onFacebookTwitterLinkedInRedditWechat Abstract In this study, a new high step-up high step-down bidirectional dc–dc converter is proposed. In the proposed converter, the voltage conversion ratio is increased by using two coupled inductors and switched capacitor circuit in both boost and buck operations. The proposed bidirectional converter in comparison with other presented high-voltage gain boost converters with two coupled inductors in the literature has the highest voltage conversion ratio. In this study, the proposed topology is analysed in all operating modes and the current and voltage stresses of all switches, voltage gain and maximum and minimum current through the inductors are calculated for both boost and buck operations. Finally, the accuracy of the obtained analytical results and correct operation of the proposed converter are verified through PSCAD/EMTDC simulation and experimental results. 1 Introduction In recent years, bidirectional dc–dc converters have taken more attention for their usage in fuel cell, photovoltaic system, plug-in hybrid electric vehicles [1, 2]. In these systems, the used battery charges and discharges for all the time and it is necessary to have bidirectional power flow between input source and battery [3]. Moreover, in the photovoltaic system the output voltage is low. Therefore, it should be increased by using a high-voltage gain converter [3]. In [4] a multi-input multi-output boost converter for using in photovoltaic and fuel cell systems has been presented. In this converter, the input current is continuous and the voltage gain is increased by the use of switched capacitor circuit. This converter has unidirectional power flow. In [5, 6] two non-isolated high-voltage gain boost dc–dc converters have been presented. These converters have low voltage stresses on switches. These converters use n stages of diode–capacitor–inductor units. These converters have unidirectional power flow. In [7-13] several interleaved high-voltage gain boost dc–dc converters have been presented. The interleaved converters decrease the input current ripple. The presented boost converters in [7-9] use three-winding coupled inductors to increase voltage gain. In the presented converters in [7, 9], the voltage multiplier cells consisting diode-capacitor circuits is used. Both of these converters have the same voltage gain. The presented converter in [8] by the use of lower number of components can achieve lower voltage gain than the presented converters in [7, 9]. The presented converter in [10] uses one two-winding coupled inductor and diode-capacitor circuit to increase voltage gain. This converter is a kind of three-phase converter and uses three switches. The presented converter in [11] has the same voltage gain as [10] and uses two two-winding coupled inductors and diode capacitor circuit. The presented converter in [12] uses two three-winding coupled inductors and diode-capacitor circuit to increase voltage gain. The voltage stresses on switches in this converter are low. The presented two-phase converter in [13] uses only diode-capacitor circuit to increase voltage gain. The voltage gain of this converter can be increased by increasing the number of voltage multiplier cells. The voltage gain of this converter is lower than the presented converters in [7-12]. The presented interleaved converters in [7-13] achieve high-voltage gain for duty cycle >. In [14], a high-voltage gain converter has been presented. In this converter by using one three-winding coupled inductor and diode capacitor circuit, voltage gain is increased. All of presented converters in [7-14] are unidirectional converters. The bidirectional dc–dc converters are divided into two groups of isolated and non-isolated converters that non-isolated converters have losses lower than isolated converters [15, 16]. In [16-20], several bidirectional dc–dc converters with high-voltage conversion ratio for boost and buck operations have been presented. The presented converters in [16, 17] use switched capacitor circuits to increase voltage gain. These converters all have the same voltage gain. The presented bidirectional converters in [18, 19] provide high-voltage gain by the use of one two-winding coupled inductor. The voltage gains of these converters are lower than the presented converters in [16, 17]. The switches of presented converter in [19] by the use of an auxiliary circuit operate under soft switching. The presented bidirectional converter in [20] uses one two-winding coupled inductor, and switched capacitor circuits that include five switches and three capacitors. The presented bidirectional converter in [20] by the use of higher number of components provides the highest voltage gain in comparison with the presented bidirectional converters in [16-19]. In this paper, a new high-voltage-gain bidirectional dc–dc converter is proposed. In addition, three derivative converters of the proposed topology are proposed. The proposed topology is analysed in all operating modes and all equations of voltage and current of components for both boost and buck operations are given. Finally, the accuracy of the proposed converter is verified through PSCAD/EMTDC simulation and experimental results. 2 Proposed converter The power circuit of the proposed converter is shown in Fig. 1a. The capacitance C is considered large enough. Two coupled inductors are used to improve the voltage gain. Each coupled inductor has the magnetising inductors and , and leakage inductors and . The primary and secondary windings of each coupled inductor have and turns, respectively. Therefore, the turns ratio of the coupled inductors are considered as , and . The power circuits of the proposed converter in boost and buck operations are shown in Figs. 1b and c, respectively. Considering Figs. 1b and c, it can be seen that for using the proposed converter in boost and buck operations, only the places of input voltage source and output load are changed. In the coupled inductors, the coupling coefficient K is defined as . In other words, the ratio of inductances and , is defined as . Since the capacitor C is considered large enough, the voltage across the capacitor C can be considered as a voltage source with magnitude . Fig. 1Open in figure viewerPowerPoint Power circuit of the proposed converter (a) Proposed bidirectional dc-dc converter, (b) Power circuit in boost operation, (c) Power circuit in buck operation 2.1 Boost operation The voltage and current waveforms of the proposed converter in boost operation for ideal coupled inductors ( and ) are shown in Fig. 2a. Moreover, the equivalent circuits during one switching period () are shown in Figs. 3a and b. As shown in Fig. 2a before the moment , the switches and are turned on and the currents of inductances and are linearly decreased and reached to their minimum values and at the moment . Fig. 2Open in figure viewerPowerPoint Key waveforms (a) In boost operation, (b) In buck operation Fig. 3Open in figure viewerPowerPoint Equivalent circuits during one switching period (a) Mode 1 of boost operation, (b) Mode 2 of boost operation, (c) Mode 1 of buck operation, (d) Mode 2 of buck operation 2.1.1 First operating mode () The equivalent circuit of the first operating mode is shown in Fig. 3a. By applying Kirchhoff's Voltage Law (KVL) and Kirchhoff's Current Law in Fig. 3a the following equations are obtained: (1) (2) (3)By deriving from (3), and based on the voltage equations across the inductors, the following equation can be written as follows: (4)In the same way, by considering that is equal to , we have: (5)By substituting (5) into (1), it is concluded that: (6)By substituting (2) into (4), it is concluded that: (7)Considering (6) and (7), the voltages and are obtained by following equations: (8) (9)As mentioned before, the relationship between the leakage inductance, magnetising inductance and coupling coefficient in each coupled inductor can be defined as and . By considering ideal coupled inductors ( and ) the voltages and based on (8) and (9) are obtained as and , respectively. According to Fig. 3a the current and its transferred current to the primary winding of are equal to zero. Therefore, the current of switch is equal to or . As a result by considering (8) and (9), and are increased linearly as following equations: (10) (11)where and are the values of and at moment , respectively. Considering (10) and Fig. 3a, the transferred current of to primary side of transformer is equal to or . Therefore according to (3) and (11), can be written as follows: (12)The input current is equal to switch current of . Therefore by considering (10) and (11), and are obtained as follows: (13)By considering (8) and (9), the voltage stresses on switches ( and ) are calculated as: (14) (15)For ideal coupled inductors ( and ), the voltage stresses on these switches are equal to and , respectively. 2.1.2 Second operating mode () The equivalent circuit of this mode is shown in Fig. 3b. The current switch of is equal to . Therefore, its transferred current to the primary winding of is equal to or .The current of is equal to . Therefore, and can be obtained as follows: (16)By deriving from (16), and based on the relation between the currents and voltages of inductances and , the following equation can be written: (17)In the same way, the relation between the voltages and is obtained same as (17). Therefore, by applying KVL in Fig. 3b, and are obtained as following equations: (18) (19)As mentioned before, the relations and are valid. Considering (18) and (19), and decrease linearly according to the following equations: (20) (21)where and are the values of and at moment , respectively. By considering (16) and (20), it can be written that: (22)The current is equal to or . The current is equal to . Therefore, by considering (21), it is concluded that: (23)Considering (22) and (23), the input current can be obtained as follows: (24)Considering (18) and (19), the voltage stresses on switches and are calculated as follows: (25) (26)For ideal coupled inductors , the voltage stresses on these switches are obtained as , and , respectively. Inductors currents calculation: Based on the current balance law of capacitors, the average current of capacitors and C are equal to zero in the steady state. During first operating mode, the current of capacitor is equal to and during second operating modes it is obtained as or based on (16) is obtained as . Therefore, the average current of capacitor can be written as follows: (27)where and are the average values of and , respectively. By simplifying (27), it is concluded that: (28)where and are the maximum and minimum values of , respectively. Considering (10) and (23), and the current balance law for , we have: (29)where is the average value of . By considering Fig. 2a, and by replacing (28) into (29), it is obtained that: (30)where and are the maximum and minimum current values of , respectively. Considering Fig. 2a and (10) and (11), the values of and at moment can be rewritten as follows: (31) (32)According to Fig. 2a, the time interval of first mode can be written as . Considering (28) and (30)–(32), the maximum and minimum values of , , and can be written as: (33) (34) (35) (36) 2.2 Buck operation The voltage and current waveforms of the proposed converter in buck operation for ideal coupled inductors are shown in Fig. 2b. By comparing Figs. 2a and b, it is clear that the waveforms of the proposed converter in buck and boost operations are the same and only the direction of the currents of and , and switches – are changed. The equivalent circuits of buck operation are shown in Figs. 3c and d. Before the moment , the switches and are turned on, and the currents of inductances and are linearly decreased and reached to their minimum values and at the moment . 2.2.1 First operating mode () The equivalent circuit of the first mode is shown in Fig. 3b. and are obtained as (8) and (9), respectively. Therefore, the currents , , , and are calculated by: (37) (38) (39) (40)where and are the current values of and at moment , respectively. The voltage stresses on switches and are given by (14) and (15), respectively. In this mode, the input current is equal to zero. 2.2.2 Second operating mode () The equivalent circuit of this mode is shown in Fig. 3d. and are obtained as (18) and (19), respectively. Moreover, , , , , , , , and can be obtained as in following equations: (41) (42) (43) (44)where and are the values of and at moment , respectively. The voltage stresses on switches and are obtained by (25) and (26), respectively. Inductors currents calculation: Based on the current balance law for capacitors, the average current of capacitors and C are equal to zero. According to Fig. 3, in buck operation the value of is always obtained by . Therefore, during first mode by considering (37) and (39), is obtained as equal to . In the same way, during the second mode according to (43) and (44) the current is obtained as . Therefore, the average current of capacitor can be written as follows: (45)where , and are the average values of , and , respectively. Considering (29), (37), (44), (45) and the current balance law for capacitor , we have: (46) (47)where and are the minimum and maximum values of , respectively. Moreover, and are the minimum and maximum values of , respectively. Considering Fig. 2b and (37) and (38), and at moment can be written as (31) and (32), respectively. Therefore, by considering (31), (32), (46) and (47), the minimum and maximum values of , , and can be calculated as follows: (48) (49) (50) (51) 2.3 Voltage gain calculation According to the voltage balance law of inductors, in the steady state the average voltages of the inductors and are equal to zero. Therefore, by using the values of from (9) and (19), the following equation is written: (52)By simplifying (52), can be obtained as follows: (53)For ideal coupled inductors ( and ), is obtained as . Considering (8) and (18), we have: (54)By simplifying (54), voltage gain is written as follows: (55)The voltage gains of proposed converter in boost and buck operations are equal to and , respectively. By assuming ideal coupled inductors ( or ), the voltage gain can be simplified as follows: (56) 3 Comparison results Table 1 shows the comparison results between the proposed converter and couple inductor based converters presented in [8-12, 15]. Moreover, comparison results with presented bidirectional converters in [17-21] are given in Table 2. Tables 1 and 2 give the comparing results between the proposed converters and the presented converters in the literature from different aspects including voltage gain, voltage stresses on switches , number of switches , number of diodes , number of used multi-winding coupled inductors (two-winding or three-winding), number of typical inductors , number of capacitors and capability of bidirectional power flow. Table 1. Comparing the proposed converter with coupled inductor based converters in the literature DC–DC converters Bidirectional [8] no 2 8 two two-winding — 5 [9] no 2 6 two three-winding — 4 [10] 2 switches: 1 switch: no 3 6 one two-winding 2 3 [11] no 2 4 two two-winding — 3 [12] no 2 6 two three-winding — 5 [15] no 1 4 one three-winding — 4 proposed converter , , yes 4 — two two-winding — 1 Table 2. Comparing the proposed converter with bidirectional converters in the literature DC–DC converters Bidirectional [17] no 4 — — 2 1 [18] no 4 — — 2 1 [19] no 2 — 1 — 1 [20] no 3 — 1 2 1 [21] no 5 — 1 - 3 proposed converter , , yes 4 — 2 — 1 Considering Table 1, it can be seen that the proposed converter has the highest voltage gain comparing with other presented coupled inductor based boost. This fact is also shown in Fig. 4 for . The presented interleaved converters in [8-12] can achieve high-voltage gain only for duty cycle >. Fig. 4Open in figure viewerPowerPoint Variation of voltage gain versus duty cycle for The presented converter in [15] achieves high-voltage gain for whole range of duty cycles. The presented converters in [8-12, 15] have unidirectional power flow. The sum of numbers of switches, diodes and capacitors in the proposed converter is lower than the presented converters in [8-12, 15]. Therefore, the main advantages of the proposed converter are achieving high-voltage gain by using less number of components and capability of bidirectional power flow. Moreover, the proposed converter in comparison with the presented converters in [8-12] can achieve high-voltage gain for whole range of duty cycles. By considering Table 2, it can be seen that the proposed bidirectional converter has high-voltage gain comparing with the other presented bidirectional converters in [17-21]. This fact can be seen from Fig. 4, too. 4 Derivative converters Fig. 5 shows four derivative converters, which are derived from the proposed converter. The proposed converters on the left- and right-hand side of Fig. 5a use one coupled inductor. In the proposed converter in Fig. 5b, two three-winding coupled inductors are used to achieve the zero input current ripple and high-voltage gain. The proposed converters in Fig. 5 can be analysed similar to the main proposed converter in Fig. 1. Table 3 summarises the characteristics of converters A and B for ideal coupled inductors . Considering (14), (15), (25), (26), (56) and Table 3, it can be concluded that the voltage gain and voltage stresses on switches () for the proposed converters A and B can be obtained by choosing the turns ratio of transformers as and in the main proposed converter, respectively. Table 3. Characteristics of the proposed derivative converters A and B Characteristics Converter A Converter B , , , , , , and in boost operation , , and in buck operation Fig. 5Open in figure viewerPowerPoint Proposed derivative converters (a) Converter A (left-hand side) and converter B (right-hand side), (b) Converter C, (c) converter D, (d) Key waveforms of converter D In the proposed converter C, the turns ratios of the first coupled inductor are considered as , and . The turns ratios of the second coupled inductor are considered as , and . In this converter, by using two extra capacitors and three-winding coupled inductors the ripples of , and are eliminated. For this cause, the turns ratio of third windings of first and second coupled inductors are considered equal to 1 and the value of leakage inductors and should be considered . In this converter by assuming that the capacitors and are large enough, the voltage across the capacitors and are considered as voltage source . In this condition, all characteristics of converter C will be same as the main proposed converter shown in Fig. 1. The switches of the proposed converter in Fig. 5c operate under zero voltage switching (ZVS). The only difference between the power circuit of this converter and the main proposed converter in Fig. 1a is that this converter consists the parasitic capacitors across the power switches. Moreover, for achieving ZVS of switches in this converter, a dead time () is considered between the gate pulses of switches as shown in Fig. 5d. Fig. 5d shows the waveforms of proposed converter D for boost operation. Considering this figure, ZVS operation of switches and is always found for boost operation. Hence, only the required conditions for ZVS operation of switches and is given. Considering Fig. 5d, during time intervals of , and all of switches are turned off. On the other hand, the currents through the inductors cannot reach to zero instantaneously. As a result, during time intervals of and the parasitic capacitors of all switches should conduct. Therefore, during the capacitors and are fully discharged to zero from their initial voltages. The capacitors and are fully charged. Hence, the required time intervals for fully discharging of the capacitors and to zero value are calculated as follows: (57) (58)where and are replaced from (25) and (26) into (57) and (58). Considering Fig. 5d after the time intervals of and , the voltages across the switches and reach zero. In this condition, if the currents through the switches and have negative values, the internal diodes of these switches can be turned on. Hence, in the next mode, primarily the diodes of switches and conduct and then the pulses of these switches are applied on as shown in Fig. 5d. Therefore, ZVS operation of switches and is achieved, because at the turn on moment of this switches, the voltages across them are equal to zero. Considering Fig. 5d and (34), ZVS condition of switch is written as follows: (59)Considering Fig. 5d, (34) and (36), ZVS condition of switch is written as follows: (60)Considering , , , (59) and (60), the required conditions for ZVS of switches and in boost operation are simplified as follows: (61) (62)In the same way, in the buck operation the ZVS operation of switches and are always found because the internal diodes of these switches conduct for long time. Considering the waveforms of and from Fig. 2, the switches and can turn off at ZVS state under the conditions as and , respectively. Hence, considering , , , (49) and (51), the required conditions for ZVS turning off of switches and in buck operation is written as following inequalities: (63) (64)The voltage waveforms of proposed converter D in buck operation are same as boost operation. 5 Design consideration To verify that the inductor's currents for both buck and boost operations were in one direction, the average current of magnetising inductances ( and ) should be higher than the half of current ripple of these inductances. In boost operation for ideal coupled inductors , the value of by considering (30), (35), (36) and Fig. 2a can be written as follows: (65)The output current in the boost operation is defined as , where is the output resistance in boost operation. The value of voltage gain of boost operation is obtained from (56). By simplifying (65), the value of is obtained as follows: (66)In the same way by considering (28), (33) and (34), the value of is calculated as follows: (67)Similarly, the value of inductances in buck operations for ideal coupled inductors by considering (46)–(51) are obtained as follows: (68) (69)The average value of output current in buck operation is defined as , where is the output resistance in buck operation. By considering (10) and (28), the average current of capacitor C can be rewritten as following equation: (70)In the above equation, the negative sign shows that the voltage ripple across the capacitor C is a negative value. By simplifying (70), the size of capacitor C in boost operation is obtained as follows: (71)where is defined as for boost operation. In buck operation by considering (37) and (46), the average current of capacitor C during first operating mode can be rewritten as follows: (72)In above equation, is defined as for buck operation. Therefore, the size of capacitor C in buck operation for maximum value of can be obtained as follows: (73) 6 Efficiency calculation In this part, the conduction and switching losses of proposed converter are calculated to obtain the efficiency for both buck and boost operations. Hence, the internal resistors of diodes , switches , inductors , capacitor , forward drop voltage of diodes and forward drop voltage of switches are considered for calculating power losses. Referring Fig. 2a and (28), (30) the average values of currents of switches and diodes in boost operation are calculated as follows: (74) (75) (76)Referring to [21] and (66)–(68), conduction losses of switches and diodes for boost operation are calculated as following equations: (77) (78) (79) (80)Referring to [21, 22], (14), (15), (25), (26), (28), (30) and Fig. 2, the switching losses for the switches and are calculated as follows: (81) (82)As mentioned earlier, in boost operation the internal diodes of switches and ( and ) conduct. At the turn-on moment, the diodes and can be considered as an ideal switch because they turn on rapidly in comparison with the transients in the power circuit [21]. Hence, turn-on power loss of diodes and is neglected . Although, at turn-off, the diode current reverses for a reverse recovery time [22]. is divided into two time intervals . In , is still zero; however, in , changes from zero to its voltage stress at turn off state, linearly [22]. Hence, the turn-off power loss of diode is written as follows: (83)In the same way, the turn-off power loss of diode is calculated as follows: (84)Considering (77)–(84), the power loss of switches and diodes (switching and conduction loss) in boost operation is calculated as follows: (85) (86)The conduction losses of the capacitor C, considering (10), (23), (28) is calculated as follows: (87)The conduction losses of the inductors and , considering (10), (23), (28) and (30) is calculated as following equations: (88)The power losses calculations of the semiconductors and inductors have been presented in [21-23]. Referring to [23], in the proposed converter, the core power losses density related to the coupled inductors is calculated as follows: (89)The power loss density is in unit . , and k are called Steinmetz parameters which are usually provided by manufacturers for different core materials [23]. Typical values of can vary from 1 to 2 for ferrite materials . According to, Faraday's Law [16], it can be written that: (90)where is the core area that have been presented by manufactures for different kinds of magnetic cores [24]. N is the turns of inductor's windings. Hence, considering (90) and the waveforms of and from Fig. 2, the peak flux density of for first and second coupled inductors is obtained as follows: (91) (92)The total core loss for each inductor is equal to . where is the volume of inductor core from datasheet. Hence, considering , and (89) the total core losses of inductors is calculated as follows: (93)As a result, considering (85)–(89), the total power losses for boost operation is written as follows: (94)where is output current from Fig. 1b. The efficiency of proposed converter for boost operation , considering (94) is calculated as follows: (95)where is output power for boost operation that is written as . Similarly, for buck operation the power loss is obtained as follows: (96)where is output current from Fig. 1c. is replaced from (56) into (96). In buck operation the internal diodes of switches and ( and ) conduct. Hence, turn-on power loss of diodes and is ignored . The efficiency of proposed converter for buck operation , considering (96) is calculated as follows: (97)where is output power for buck operation that is written as . The efficiency of proposed converter for boost and buck operations versus input voltage and output power are plotted in Figs. 6a and b. Hence, referring to (94)–(97), the analytical efficiency of proposed converter can be calculated for input voltages , , and for boost and buck operations. Moreover, the efficiency curve is obtained for one input voltage in the both operations by using simulation results. The used parameters are as , and . By considering the used power MOSFET as IRF740, and the used diodes as MUR1560G, the other parameters are as , , , , , , , and . Fig. 6Open in figure viewerPowerPoint Efficiency of proposed converter (a) Boost operation, (b) Buck operation 7 Simulation and experimental results In this section, in order to verify the obtained theoretical analysis the experimental results are extracted for both boost and buck operations. The used experimental parameters are summarised in Table 4. The simulation results for verifying ZVS of switches in boost operation of proposed derivative converter D are used. The simulation parameters of the converter D are summarised in Table 5. Table 4. Experimental parameters for boost and buck operations , , , Table 5. Simulation parameters for proposed converter D , , , 7.1 Experimental results In the boost operation of proposed converter, the values of input voltage, output resistance and the output capacitor resistance are considered as , and , respectively. Fig. 7 shows the experimental results for proposed converter in boost operation. Fig. 7Open in figure viewerPowerPoint Experimental results for boost operation (a) (top-right-hand side), (top-left-hand side), (bottom-right-hand side) and (bottom-left-hand side), (b) (top-right-hand side), (top-left-hand side), (bottom-right-hand side) and (bottom-left-hand side), (c) (left-hand side) and (right-hand side) According to (55) the value of output voltage for the determined parameters in Table 4, is obtained as that can be verified by left figure of Fig. 7c. The voltage for the selected parameters in Table 4 can be calculated from (53) as that can be verified by using right figure of Fig. 7c. Hence, the voltage stresses on switches , , and from (25), (26), (14) and (15) are calculated as , , and , respectively. The calculated values for voltage stresses on switches can be verified from Figs. 7a and b right-hand side. Referring to (33)–(36) and considering Fig. 2a, the maximum and minimum currents through the switches can be calculated as , , , , , , and . These calculations can be verified by experimental results in Figs. 7a and b left-hand side. The directions of the currents consistent with the assumed directions in Fig. 1. The negative calculated values for currents and show that the internal diodes of switches and are conducting in boost operation (their MOSFETs are turned off). The positive calculated values for currents and verify that MOSFETs of switches and are conducting. In this condition, referring to (94) and (95), the calculated efficiency is equal to . In the buck operation, the values of input voltage, output resistance and the output capacitor resistance are considered as , and , respectively. Fig. 8a shows the experimental results for the proposed converter in buck operation. The output voltage is shown in Fig. 8a that is verified by (55) . The voltage stresses on switches in buck operation are same as boost operation that are shown in Figs. 7a and b. Same as boost operation, the voltage for the selected parameters in Table 4 can be calculated from (53) as . Referring to (48)–(51) and considering Fig. 2b, the maximum and minimum currents through the switches can be calculated as , , , , , , and . These calculated values can be verified by experimental results in Fig. 8a. The current directions of the currents , , and consistent with the assumed directions in Fig. 1. The positive calculated values for currents and show that the MOSFETs of switches and are conducting in buck operation. The negative calculated values for currents and verify that the internal diodes of switches and are conducting. The experimental results verify these facts. In this condition, referring to (94) and (95), the calculated efficiency is equal to . Fig. 8Open in figure viewerPowerPoint Voltage stresses on switches in buck operation are same as boost operation (a) Experimental results for buck operation, (b) Simulation results for boost operation of proposed converter D, (c) Experimental prototype of the proposed converter 7.2 Simulation results As explained in Section 4, the switches of the proposed converter D in Fig. 5c operate under ZVS condition. Fig. 8b verify this fact. Considering the waveforms of , , and in Fig. 8b, it can be seen that during the dead time the voltages across the switches and reached zero. In this condition, the currents and have negative values, hence, at first the internal diodes of these switches are turned on and then by applying the pulse gates of switches ( and ), the switches and are turned on under ZVS condition. ZVS operation of switches and is always found in boost operation. According to (55) the value of output voltage for the determined parameters in Table 5, is obtained as that can be verified by Fig. 8b. The voltage for the selected parameters in Table 5, can be calculated from (53) as . Hence, the voltage stresses on switches , , and from (25), (26), (14) and (15) are calculated as , , and , respectively. Therefore, the calculated voltage stresses on switches are verified by Fig. 8b. The power circuit of experimental prototype of proposed converter is shown in Fig. 8c. 8 Conclusion In this paper, a new bidirectional dc–dc converter was proposed. The main advantage of the proposed converter comparing with the other presented coupled inductor based converters in the literature is highest voltage conversion ratio in both buck and boost operations by the use of lower number of components, respectively. In this paper, three derivative converters based on the proposed topology were proposed. The proposed topology was analysed in all operating modes and the values of current stresses of all switches, voltage stresses on all switches and voltage gain was calculated for both boost and buck operation. Finally, all that were given theoretical were reconfirmed by using experimental and PSCAD/EMTDC simulation results for boost and buck operations. 9 References 1Babaei E., Saadatizadeh Z., and Laali S.: ‘A new topology of bidirectional buck-boost dc/dc converter with capability of soft switching and input current ripple cancellation’, Iran. J. Electr. Electron. Eng., 2016, 12, (2), pp. 134– 146 2Tang Y., and Khaligh A.: ‘Bidirectional resonant dc-dc step-up converters for driving high-voltage actuators in mobile microrobots’, IEEE Trans. Power Electron., 2016, 31, (1), pp. 340– 352 3Zhang J.: ‘ Bidirectional dc-dc power converter design optimization, modeling and control’. PhD. Thesis, Virginia Polytechnic Institute and State University, 2008 4Babaei E., and Abbasi O.: ‘Structure for multi-input multi-output dc–dc boost converter’, IET Power Electron., 2016, 9, (1), pp. 9– 19 5Nouri T., Hosseini S.H., and Babaei E.: ‘Analysis of voltage and current stresses of a generalized step-up dc-dc converter’, IET Power Electron., 2014, 7, (6), pp. 1347– 1361 6Nouri T., Hosseini S.H., and Babaei E. et al.: ‘Generalised transformerless ultra step-up dc–dc converter with reduced voltage stress on semiconductors’, IET Power Electron., 2014, 7, (11), pp. 2791– 2805 7Li W., Zhao Y., and Wu J. et al.: ‘Interleaved high step-up converter with winding-cross-coupled inductors and voltage multiplier cells’, IEEE Trans. Power Electron., 2012, 27, (1), pp. 133– 143 8Babaei E., Saadatizadeh Z., and Mohammadi ivatloo B.: ‘A new interleaved bidirectional zero voltage switching dc/dc converter with high conversion ratio’, J. Circuits, Syst. Comput., 2017, 26, (6), pp. 1– 25 9He L., and Liao Y.: ‘An advanced current-auto-balance high-step-up converter with a multi-coupled inductor and voltage multiplier for a renewable power generation system’, IEEE Trans. Power Electron., 2016, 31, (10), pp. 133– 143 10Revathi B S., and Prabhakar M.: ‘Transformerless high-gain dc–dc converter for microgrids’, IET Power Electron., 2016, 9, (6), pp. 1170– 1179 11Hu X., Dai G., and Wang L. et al.: ‘A three-state switching boost converter mixed with magnetic coupling and voltage multiplier techniques for high gain conversion’, IEEE Trans. Power Electron., 2016, 31, (4), pp. 2991– 3001 12Tseng K.C., Chen J.Z., and Lin J.T. et al.: ‘High step-up interleaved forward-flyback boost converter with three-winding coupled inductors’, IEEE Trans. Power Electron., 2015, 30, (9), pp. 4696– 4703 13Kishore Prabhala V.A., Fajri P., and Prasad Gouribhatla V.S. et al.: ‘A dc–dc converter with high voltage gain and two input boost stages’, IEEE Trans. Power Electron., 2016, 31, (6), pp. 4206– 4215 14Tseng Ch., Lin J.T., and Huang Ch.Ch.: ‘High step-up converter with three-winding coupled inductor for fuel cell energy source applications’, IEEE Trans. Power Electron., 2015, 30, (2), pp. 574– 581 15Yao C., Ruan X., and Wang X. et al.: ‘Isolated buck-boost dc/dc converters suitable for wide input-voltage range’, IEEE Trans. Power Electron., 2011, 26, (9), pp. 2599– 2613 16Ardi H., Ajami A., and Kardan F. et al.: ‘Analysis and implementation of a non-isolated bidirectional DC-DC converter with high voltage gain’, IEEE Trans. Power Electron., 2016, 63, (8), pp. 4878– 4888 17Ardi H., Ahrabi R.R., and Najafi Ravadanegh S.: ‘Analysis and implementation of a non-isolated bidirectional dc-dc converter with high voltage gain’, IEEE Trans. Power Electron., 2014, 7, (12), pp. 3033– 3044 18Narasimharaju B.L., Dubey S.P., and Singh S.P.: ‘Design and analysis of coupled inductor bidirectional dc-dc convertor for high-voltage diversity applications’, IET Power Electron., 2012, 5, (7), pp. 998– 1007 19Das P., Mousavi S.A., and Moschopoulos G.: ‘Analysis and design of a non-isolated bidirectional ZVS-PWM dc–dc converter with coupled inductors’, IEEE Trans. Power Electron., 2010, 25, (10), pp. 2630– 2641 20Hsieh Y.P., Chen J.F., and Yang L.Sh. et al.: ‘High-conversion-ratio bidirectional dc–dc converter with coupled inductor’, IEEE Trans. Power Electron., 2014, 61, (1), pp. 210– 222 21Babaei E., Shokati Asl E., and Hasan Babayi M. et al.: ‘Developed embedded switched-Z-source inverter’, IET Power Electron., 2016, 9, (9), pp. 1828– 1841 22Babaei E., and Shokati Asl E.: ‘A new topology for Z-source half-bridge inverter with low voltage stress on capacitors’, Electr. Power Syst. Res., 2016, 140, pp. 722– 734 23Czogalla J., Li J., and Sullivan Ch.R.: ‘ Automotive application of multi-Phase coupled-inductor dc-dc converter’. Industry Applications Conf., Salt Lake City, USA, 2003 24https://coefs.uncc.edu/…/Transformer-and-Inductor-Design-Handbook_Chapter_3.pdf Citing Literature Volume10, Issue12October 2017Pages 1556-1571 FiguresReferencesRelatedInformation
Referência(s)