Emphasis on switch selection and its switching loss comparison for on‐board electric vehicle charger
2019; Institution of Engineering and Technology; Volume: 12; Issue: 6 Linguagem: Inglês
10.1049/iet-pel.2018.6070
ISSN1755-4543
AutoresManaswi Srivastava, Pavan Singh Tomar, Arun Kumar Verma,
Tópico(s)Wireless Power Transfer Systems
ResumoIET Power ElectronicsVolume 12, Issue 6 p. 1385-1392 Research ArticleFree Access Emphasis on switch selection and its switching loss comparison for on-board electric vehicle charger Manaswi Srivastava, Corresponding Author Manaswi Srivastava 2016ree9506@mnit.ac.in Department of Electrical Engineering, Malaviya National Institute of Technology, Jaipur, IndiaSearch for more papers by this authorPavan Singh Tomar, Pavan Singh Tomar Department of Electrical Engineering, Malaviya National Institute of Technology, Jaipur, IndiaSearch for more papers by this authorArun Kumar Verma, Arun Kumar Verma Department of Electrical Engineering, Malaviya National Institute of Technology, Jaipur, IndiaSearch for more papers by this author Manaswi Srivastava, Corresponding Author Manaswi Srivastava 2016ree9506@mnit.ac.in Department of Electrical Engineering, Malaviya National Institute of Technology, Jaipur, IndiaSearch for more papers by this authorPavan Singh Tomar, Pavan Singh Tomar Department of Electrical Engineering, Malaviya National Institute of Technology, Jaipur, IndiaSearch for more papers by this authorArun Kumar Verma, Arun Kumar Verma Department of Electrical Engineering, Malaviya National Institute of Technology, Jaipur, IndiaSearch for more papers by this author First published: 01 May 2019 https://doi.org/10.1049/iet-pel.2018.6070Citations: 8AboutSectionsPDF ToolsRequest permissionExport citationAdd to favoritesTrack citation ShareShare Give accessShare full text accessShare full-text accessPlease review our Terms and Conditions of Use and check box below to share full-text version of article.I have read and accept the Wiley Online Library Terms and Conditions of UseShareable LinkUse the link below to share a full-text version of this article with your friends and colleagues. Learn more.Copy URL Share a linkShare onFacebookTwitterLinkedInRedditWechat Abstract A DC/DC converter plays a vital role in on-board electric vehicle (EV) charger. Among the many converter topologies, the phase-shifted full-bridge DC/DC converter (PSFBDC) is widely used. For improved efficiency, PSFBDC is embedded with passive auxiliary networks resulting in several unique configurations for achieving soft-switching in entire load range. The architecture of the chosen DC/DC converter configuration should comply with the society of automotive engineers (SAE) standards. Although, the conduction loss of the converter can be minimised by choosing a proper topology configuration and proper switching of the configuration, the selection of appropriate for PSFBDC plays an important role in enhancing the overall efficiency. An optimistic method for the selection of switch configuration satisfying the reduced power loss and cost-effective aspects is presented. For demonstrating the proposed method, a 1 kW 100 kHz system with commercially available switches IRFP460, IXFN64N60P, and IPP65R045C7 is chosen and the obtained results are presented. 1 Introduction With the depletion of fossil fuel and increase in awareness of using renewable energy resources, electric vehicle (EV)-based transportation has become a major focus. In general, a power electronics system (PES) interfacing the utility grid and battery pack of EV plays a key role in enhancing the efficient utilisation of EV. Mainly, PES for EV is of two types: off-board/EV service equipment (EVSE) and on-board which may be installed in a home garage, parking lot of commercial offices, along a street etc for charging the EV battery. The generalised block diagram of on-board and off-board charging is shown in Fig. 1, with emphasis on the internal input architecture of on-board charger. The internal architecture comprises of two stages: AC/DC and DC/DC. The AC stage is responsible for improving the input power factor, while the DC stage provides the suitable voltage and current in accordance to the battery profile [1]. Nowadays, the development of highly efficient and economic EV battery chargers is of high interest. For fulfilling these emerging features of EV battery charging schemes, battery management system (BMS) is essential in which state-of-charge (SOC) estimation and state-of-health (SOH) monitoring are of particular importance. To harness these competing objectives, including safety, lifetime, charging time, and a health-aware fast charging strategy is given in [2, 3]. Fig. 1Open in figure viewerPowerPoint AC/DC and DC/DC charging description and a single-phase input architecture of an on-board charger The society of automotive engineers (SAE) have drafted the standard charging levels for on-board and off-board chargers as shown in Fig. 2 [4]. According to SAE standards, AC charger ratings should be . In general, the phase-shifted full-bridge DC/DC converter (PSFBDC) are widely accepted for medium power range applications owing to its features like simple structure, clamping of voltage stress across the switch to the DC level, constant frequency, high power density, high efficiency, and less electromagnetic interference [5-7]. Hence, with these attractive features, PSFBDC qualifies as one of the most suitable configuration for the on-board charger. Fig. 2Open in figure viewerPowerPoint SAE standards for AC charging and DC charging of EVs In practice, soft switching is adopted for reducing the switch stress and power losses to enhance charger life and efficiency [8]. To achieve this, an auxiliary network is included in the full-bridge DC/DC converters (FBDC) for supporting the conventional FBDC configuration to the main zero voltage switching (ZVS) range with the full-load variation [9, 10]. These auxiliary networks are generally classified as: series auxiliary and parallel auxiliary networks [11, 12]. The series auxiliary networks results in excessive duty cycle loss and hazardous parasitic ringing. Thus, the ZVS is lost during light load condition. However, the use of parallel auxiliary network leads to higher conduction loss because of the increased current through the switches which reduce the efficiency of the converter [13, 14]. Nevertheless, parallel auxiliary network is a trade-off between the duty cycle loss and ZVS range. Therefore, the parallel auxiliary networks have more advantages over series auxiliary network because its adaptation are totally based on the remedies of the series auxiliary network [15]. The qualitative analysis of a full-bridge DC/DC converter with auxiliary network has been done in [16]. However, it lacks of quantitative analysis. In [17], the derivation of primary current of conventional PSFBDC along with the calculation of switching power loss has been carried out. Nevertheless, the loss analysis pertaining to other converter components has been overlooked. While the complete loss analysis is uninvestigated in [18], it has been partially carried out in [19] for leading and lagging leg. Recapitulating the above works, the following observations are noteworthy: The available chargers are designed for a particular SAE charging levels with some quantitative and qualitative analysis. A detailed quantitative analysis of the converter current for a chosen SAE level remains unattained. On this line, the paper attempts to provide a methodology for choosing an appropriate switch among the various commercially available switches. For demonstrating the proposed approach, the popular -type auxiliary network-based PSFBDC (refer Fig. 3) with ZVS capability for the entire load range is considered [20]. The proposed method is elucidated considering a power rating of 1 KW and 100 kHz switching frequency. The commercial switches IRFP460, IXFN64N60P, and IPP65R045C7 are chosen for validating the effect of switch selection on conduction loss and efficiency of the converter. Fig. 3Open in figure viewerPowerPoint Full-bridge DC/ DC converter topology with auxiliary network The rest of the paper is organised as follows. Section 2 addresses the loss analysis model; Section 3 presents the power loss analysis. The rationale for switch selection of the converter is described in Section 4, and Section 5 presents the conclusion. 2 Loss analysis model The converter considered for analysis is a PSFBDC with low leakage inductance. The key waveform of the PSFBDC is shown in Fig. 4a, and the loss analysis waveforms of the converter is shown in Fig. 4b. The loss analysis of the converter is as follows. Fig. 4Open in figure viewerPowerPoint Topology (a) Key operating waveform, and (b) Loss analysis waveforms During freewheeling mode shown in Fig. 5a at , the voltage across points A and B , drops to zero. Hence, and simultaneously conduct. Fig. 5b shows the equivalent circuit of this mode. The primary voltage decays to when the current is flowing through and and voltage drop across and becomes . The rate of current decay is quick due to low value of . Fig. 5Open in figure viewerPowerPoint Freewheeling mode equivalent circuit (a) Topology operating in freewheeling mode, and (b) Equivalent circuit As the value of the current reaches zero, output current will be shared equally by the output rectifier diodes. The primary and secondary voltages of the main-transformer clamped to zero when and are in conduction mode. Hence, the input voltage fully applied on . Therefore, the primary current increase in the negative direction linearly as shown in Fig. 6a. Its equivalent circuit is shown in Fig. 6b. The rise of current in negative direction is very short due to low value of inductance. Hence, the primary current is represented with dotted lines in Fig. 4b. Fig. 6Open in figure viewerPowerPoint Communication mode equivalent circuit (a) Topology operating in communication mode, and (b) Equivalent circuit 3 Power loss analysis Using Section 2, the conduction losses in the switch/ MOSFET is calculated. Although, the main power loss components in the circuit are rectifier diodes, transformer, auxiliary inductor, and filter inductor, the analysis here is restricted to the conduction loss of the switches. The following assumptions have been made for facilitating the analysis. Before , ripple in primary current is zero. Dead time is neglected. Perfect soft-switching is assumed. Other component losses are keeping constant. The switch parameters are taken from data-sheets [21-23] and assumed to be non-ideal. It is to be mentioned that the calculation is carried for maximum current rating of the on-board charger configuration for the safety purpose. Hence, the calculated values shown in the upcoming sections are for 20 A current rating according to SAE standards. 3.1 MOSFET losses Three types of losses are mainly associated with the MOSFET, they are, conduction loss, switching loss, and gate driver loss. For the loss analysis, following parameters are considered [20], input voltage , output voltage , output current , switching frequency , switching time , main transformer turns ratio n = 4.67, main transformer leakage inductance , auxiliary inductors and are 250 μH and 100 μH, respectively, auxiliary capacitors and are 2.2 μF each, and filter inductor . Further, the losses associated with the power switch of the DC/DC converter are calculated. 3.1.1 MOSFET conduction loss MOSFET conduction loss is calculated from main channel and anti-parallel body diode. The main channel loss is calculated using the on-state drain-source resistance and rms current through the switch. The anti-parallel body diode conduction loss is calculated by the intrinsic diode forward voltage drop and the average diode current. For the chosen configuration, it is understood that the current flowing in leading leg and lagging leg is different. Therefore, Fig. 4b is used as the reference for the calculation. The current in switch during interval can be written as: (1)During interval (2)where is the current in switch , is output current, n is main transformer turns ratio, is current through auxiliary inductor , is duty cycle, is transformer turns ratio, and is switching time period of the MOSFET. Equations (1) and (2) correspond to the current flowing through the MOSFET and the anti-parallel body-diode of the MOSFET, respectively. Further, it can be observed from Figs. 5a and 6a that the MOSFET provides the path for the current when it is greater than zero otherwise the path is provided by its anti-parallel diode. The rms current flowing through the switch is expressed as: (3)Finally, the conduction loss of is calculated using: (4)where is the on-state resistance of whose value is 270, 96, and 45 mΩ for IRFP460, IXFN64N60P, and IPP65R045C7, respectively. Further, the average current of the MOSFET anti-parallel body diode is calculated as: (5)where is the current passing through anti-parallel diode of switch , and is the switching time of the MOSFET. Finally, the conduction loss of is calculated using: (6)where is the forward voltage drop of the anti-parallel diode of switch whose value is 1.8, 1.5, and 0.9 V for IRFP460, IXFN64N60P, and IPP65R045C7, respectively, and is the average current passing through anti-parallel diode of switch . (7)where is the total conduction loss in leading leg of the converter, is the conduction loss in switch , and is the conduction loss in switch . Similarly, the lagging leg current during interval (8)For (9)where is the instantaneous current of switch , and current from auxiliary inductor . From the plot, it is analysed that there is a time period from 0 to in which anti-parallel body diode is conducting and from to MOSFET is conducting. The is calculated by using following relation: (10) (11)where is the time period of current through switch to reach zero current, is auxiliary inductor, is the output current, and is the switching time of MOSFET. The rms current is calculated by using following formula: (12)The conduction loss for IRFP460, IXFN64N60P, and IPP65R045C7 is calculated by: (13)The average current of the lagging leg MOSFET anti-parallel body diode can be expressed as: (14)Finally, the conduction loss of the MOSFET anti-parallel body diode can be calculated as: (15)where is the power loss in anti-parallel diode of switch , is voltage across anti-parallel diode of switch , and is the average current passing through anti-parallel diode of switch . Thus, the conduction loss of the lagging leg can be expresses as: (16)where is the total conduction loss in lagging leg, conduction loss in switch , and is the conduction loss in anti-parallel diode of switch . 3.1.2 MOSFET switching loss The operating range of the converter is dependent on the turn-on and turn-off time of the converter switch. Figs. 7a and b show the characteristics of the MOSFET. It is to be noted that for ZVS, the turn-on losses of the converter is almost zero. Hence, it is ensured that the losses will occur during turn-off period. The mathematical expression during turn-off loss of the MOSFET is given as: (17)where is the power loss during turn-off of the switch, is the input voltage, is the drain current at time , and is the switching time of the switch. Fig. 7Open in figure viewerPowerPoint MOSFET waveform in (a) Turn-on, and (b) Turn-off It is observed from (17) that the turn-off loss of the switch is directly proportional to the time period of the switch during turn-off state. The key factor that affects time period from to is the magnitude of the driving current at the plateau region. Hence, the switching loss for the leading leg and lagging leg (Fig. 8) can be calculated as: (18)where is power loss in switch during turn-off, is the input voltage, is the current flowing through at , and is the switching time. (19)where is power loss in switch during turn-off, is the current flowing through at , is the input voltage, and is the switching time. Fig. 8Open in figure viewerPowerPoint MOSFET current waveform in (a) leading leg, and (b) lagging leg 3.1.3 Gate driver loss The driving loss are totally dependent on the charging and discharging of the input capacitor of the MOSFET is expresses as: (20)where is the power loss in the gate driver, is the gate capacitance, is the driver voltage, is the switching time period, and is the total gate charge. In (20) is the gate capacitance and is the total gate charge found in the data sheet and is the driver voltage. The relationship between and of the MOSFET is shown in Fig. 9. Where is the total gate charge and is the gate to source voltage of the MOSFET. Fig. 9Open in figure viewerPowerPoint MOSFET and relationship 4 Switch selection First, IRFP460, IXFN64N60P, and IPP65R045C7 conduction loss are calculated. Then, the switch with lower conduction loss is opted for PSFBDC. The specifications of the basic PSFBDC are given in Table 1 and the specifications of the switches used are given in Table 2. Table 1. Parameters used in converter design Parameter Symbol Ratings/Parameters input voltage 400 V output voltage 54 V output current 20 A switching frequency 100 kHz switching time 10 μs transformer ratio 14:3 leakage inductance 6 μH auxiliary inductor 250 μH auxiliary inductor 100 μH auxiliary capacitor 2.2 μF filter inductance 27 μH MOSFET IRFP460 IXFN64N60P IPP65R045C7 rectifier diodes MUR3040 Table 2. Specifications from data-sheet [21-23] Parameter Description IRFP460 IXFN64N60P IPP65R045C7 drain to source voltage 500 V 600 V 700 V MOSFET drain current at 25 °C 20 A 50 A 46 A MOSFET on-state resistance 270 m 96 m 45 m body diode reverse recovery time 860 ns 200 ns 725 ns power conduction at 25 °C. 280 W 700 W 227 W Equations (3), (5), (9), (11), and (15) are used for the calculation of leading leg conduction loss and (23), (25), (29), (31), and (35) are used for calculating the lagging leg conduction loss for IRFP460, IXFN64N60P, and IPP65R045C7, respectively, and then comparison is shown in Table 3 according to SAE standards for AC-level charging. It is observed that the total conduction loss in IXFN64N60P is reduced to half of the losses that has occurred in conventionally used IRFP460 and in IPP65R045C7 it is reduced up to four times in comparison with IRFP460. Fig. 10 shows the total power percentage loss of the switch configuration (SC): SC1, SC2, and SC3, where SC1 is IRFP460, SC2 is IXFN64N60P, and SC3 is IPP65R045C7 and it is found that the SC1 i.e. IRFP460 having massive power loss at the higher rating of the system, whereas SC2 having nominal switch conduction loss and SC3 having minimal switch conduction loss associated with the converter configuration. Table 3. Power loss associated with switches with SAE standards Power dissipation in watts, W SAE SC PQ1 PD1 PQ4 PD4 Plead. Plag. PT 120 V, 12 A SC1 2.85 0.22 0.91 0.13 6.14 2.08 8.22 SC2 1.01 0.18 0.33 0.11 2.38 0.88 3.26 SC3 0.47 0.11 0.15 0.07 1.16 0.44 1.6 120 V, 15 A SC1 3.62 0.22 0.91 0.09 7.68 2 9.68 SC2 1.29 0.18 0.33 0.08 2.94 0.82 3.76 SC3 0.6 0.11 0.15 0.05 1.42 0.4 1.82 240 V, 16 A SC1 3.78 0.22 0.91 0.08 8 1.98 9.98 SC2 1.34 0.18 0.33 0.07 3.04 0.8 3.84 SC3 0.63 0.11 0.15 0.04 1.48 0.38 1.86 240 V, 20 A SC1 7.02 0.22 0.91 0.02 13.62 1.86 16.34 SC2 3.65 0.18 0.33 0.02 5.04 0.7 8.36 SC3 1.99 0.11 0.15 0.01 2.42 0.32 4.52 Fig. 10Open in figure viewerPowerPoint Calculated efficiency of the converter with different switches The efficiency behaviour of the PSFBDC with IRFP460, IFXN64N60P, and IPP65R045C7 is shown in Fig. 11 and its comparison is presented in Table 4. It is analysed that for 1 kW system the efficiency of PSFBDC by using IXFN64N60P is increased by 1%, where as in IPP65R045C7 it is increased by 1.37% w.r.t. the efficiency calculated using IRFP460. It is noted that, the analysis consider the efficiency at the rated value of the PSFBDC. Table 4. Efficiency comparison of different switches with increase in power Output power, W IRFP 460 IXFN64N60P IPP65R045C7 100 64.30 67.78 69.59 200 78.27 80.79 82.07 300 84.38 86.32 87.29 400 87.81 89.38 90.15 500 90.00 91.32 91.97 600 91.53 92.66 93.21 700 92.65 93.64 94.13 800 93.51 94.39 94.82 900 94.19 94.98 95.37 1000 94.74 95.46 96.11 Fig. 11Open in figure viewerPowerPoint Calculated efficiency of the converter with different switches It is to be noted that, this improvement in the efficiency of the converter emerges only by considering conduction loss parameter at the primary side of the topology and other parameters of the converter remains untouched even the secondary-side power diodes, MUR3040. It makes a considerable improvement in the efficiency of the converter. Table 5 shows the cost comparison between the SCs. Fig. 12 shows percentage variation of the cost of the SCs, and it is observed that the cost of IXFN64N60P is almost six times the IRFP460 and almost two times of IPP65R045C7. Hence, from Tables 4 and 5, it is observed that the IPP65R045C7 is the favourable switch configuration for the full-bridge DC/DC converter with auxiliary circuit then IRFP460 and IXFN64N60P. Although, the cost of IPP65R045C7 is almost three times higher than IRFP460 and almost half the cost of IXFN64N60P. The advantage of using IPP65R045C7 over IRFP460 is its efficiency which is increased by 1.37% at full-load condition and its high durability than IFRP460. In most practical cases, the IRFP460 is not durable due to its higher switching losses (high ). In comparison withXFN64N60P, it is cheaper and its efficiency is increased by 1% with reduced losses. It is noted that the price of the switch is highly market dependent. Table 5. Cost comparison of selected switches [21-23] Cost IRFP460 IXFN64N60P IPP65R045C7 INR 295.31 1718.7 820.86 USD 4.16 24.19 11.55 Fig. 12Open in figure viewerPowerPoint Cost comparison of IRFP460, IXFN64N60P, and IPP65R045C7 5 Conclusion A complete loss analysis for selecting an appropriate switch for the PSFBDC has been carried out mathematically for resulting in improved efficiency of the configuration is presented only by considering a single parameter that is conduction loss. Among the chosen commercial switches. It is found that IPP65R045C7 is the suitable MOSFET configuration with higher efficiency, low switching loss, and optimal cost in comparison with IRFP460 and IXFN64N60P. The overall performance of the system is expected to improve by such considerations. The work here would be helpful in further improving the efficiency of the PSFBDC for EV charging. 6 Acknowledgment The authors gratefully acknowledge the grant given by SERB-DST India with grant no. ECR/2016/001920 to carry out this research work. 7 References 1Gong X., and Rangaraju J.: ' Taking charge of electric vehicles-both in the vehicle and on the grid' ( Texas Instruments, Dallas, TX, USA, 2018), pp. 1– 13 2Hu X., Yuan H., and Zou C. et al.: 'Co-estimation of state of charge and state of health for lithium-Ion batteries based on fractional-order calculus', IEEE Trans. Vehicular Technology, 2018, 67, (11), pp. 10319– 10329 3Zou C., Hu X., and Wei Z. et al.: 'Electrochemical estimation and control for lithium-Ion battery health-aware fast charging', IEEE Trans. Ind. Electron., 2018, 65, (8), pp. 6635– 6645 4Williamson S.S., Rathore A.K., and Musavi F.: 'Industrial electronics for electric transportation: current state-of-the-art and future challenges', IEEE Trans. Ind. Electron., 2015, 62, (5), pp. 3021– 3032 5McGrath B.P., Holmes D.G., and McGoldrick P.J. et al.: 'Design of a soft-switched 6-kW battery charger for traction applications', IEEE Trans. Power Electron., 2007, 22, (4), pp. 1136– 1144 6Jiang Y., Chen Z., and Pan J.: 'Zero-voltage switching phase shift full-bridge step-up converter with integrated magnetic structure', IEE Proc. Power Electron., 2010, 3, (5), pp. 732– 739 7Safaee A., Jain P., and Bakhshai A.: 'A ZVS pulsewidth modulation full-bridge converter with a low-RMS-current resonant auxiliary circuit', IEEE Trans. Power Electron., 2016, 31, (6), pp. 4031– 4047 8Hsieh Y.C., and Huang C.S.: 'Li-ion battery charger based on digitally controlled phase-shifted full-bridge converter', IEE Proc. Power Electron., 2011, 4, (2), pp. 242– 247 9Hua G., Lee F.C., and Jovanovic M.M.: 'An improved PWM ZVS full-bridge zero-voltage-switched PWM converter using a saturable inductor', IEEE Trans. Power Electron., 1993, 8, (4), pp. 530– 534 10Redl R., Sokal N.O., and Balogh L.: 'A novel soft-switching full bridge dc–dc converter: analysis, design considerations, at 1.5 kW, 100kHz', IEEE Trans. Power Electron., 1991, 6, (4), pp. 408– 418 11Cho J.G., Sabate J.A., and Lee F.C.: ' Novel full bridge zero-voltage transition PWM DC/DC converter for high power applications'. Proc. IEEE APEC, Orlando, FL, USA, February 1994, pp. 143– 149 12Chen Z., Ji B., and Ji F. et al.: ' A novel ZVS full-bridge converter with auxiliary circuit'. Proc. IEEE APEC, Palm Springs, CA, USA, February 2010, pp. 1448– 1453 13Pahlevaninezhad M., Drobnik J., and Jain P.K. et al.: 'A load adaptive control approach for a zero-voltage-switching DC/DC converter used for electric vehicles', IEEE Trans. Ind. Electron., 2012, 59, (2), pp. 920– 933 14Jain P.K., Kang W., and Soin H. et al.: 'Analysis and design considerations of a load and line independent zero voltage switching full bridge DC/DC converter topology', IEEE Trans. Power Electron., 2002, 17, (5), pp. 649– 657 15Chen Z., Ji B., and Ji F. et al.: ' Analysis and design considerations of an improved ZVS full-bridge DC–DC converter'. Proc. IEEE APEC, Palm Springs, CA, USA, February 2010, pp. 1471– 1476 16Kim E.S., Joe K.Y., and Kye M.H. et al.: 'A improved soft-switch PWM FB DC/DC converter for reducing conduction losses', IEEE Trans. Power Electron., 1999, 14, (2), pp. 258– 264 17Sabate J.A., Vlatkovic V., and Ridley R.B. et al.: ' Design considerations for high-voltage high-power full-bridge zero-voltage switched PWM converter'. Proc. IEEE APEC, Los Angeles, CA, USA, March 1990, pp. 275– 284 18Redl R., Balogh L., and Edwards D.W.: ' Optimum ZVS full-bridge DC/DC converter with PWM phase-shift control: analysis, design considerations, and experimental results'. Proc. IEEE APEC, Orlando, FL, USA, February 1994, pp. 159– 165 19Emami Z., Nikpendar M., and Shafiei N. et al.: ' Leading and lagging-legs power loss analysis in ZVS phase-shift full bridge converter'. Proc. PEDSTC, Tehran, February 2011, pp. 632– 637 20Chen Z., Liu S., and Shi L. et al.: 'Power loss analysis and comparison of two full-bridge converters with auxiliary networks', IET Power Electron., 2012, 5, (9), pp. 1934– 1943 21'Elements14', Available at https://in.element14.com/search?st=irfp460, IRFP460 datasheet, accessed September 2018 22'Mouser Electronics', Available at https://www.mouser.in/ProductDetail/IXYS/IXFN64N60P?qs=t7yjd2JO%2fgSWipR5myQ3Pg==, IXFN64N60P datasheet, accessed September 2018 23'Mouser Electronics', https://www.mouser.in/ProductDetail/Infineon-Technologies/IPP65R045C7?qs=INZo0Duj8JRUOhZGm7%2FLCQ%3D%3D, IPP65R045C7 datasheet, accessed September 2018 Citing Literature Volume12, Issue6May 2019Pages 1385-1392 FiguresReferencesRelatedInformation
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