Compact metamaterial inspired conformal dual‐band antenna loaded with meander lines and fractal shaped inductor for Wi‐Fi and WiMAX applications
2019; Institution of Engineering and Technology; Volume: 13; Issue: 13 Linguagem: Inglês
10.1049/iet-map.2018.6008
ISSN1751-8733
Autores Tópico(s)Metamaterials and Metasurfaces Applications
ResumoIET Microwaves, Antennas & PropagationVolume 13, Issue 13 p. 2349-2359 Research ArticleFree Access Compact metamaterial inspired conformal dual-band antenna loaded with meander lines and fractal shaped inductor for Wi-Fi and WiMAX applications Ratikanta Sahoo, Corresponding Author Ratikanta Sahoo ratikanta.sahoo15@gmail.com orcid.org/0000-0001-5050-9292 Department of Electronics and Communication Engineering, National Institute of Technology Warangal, Warangal, Telangana, 506004 IndiaSearch for more papers by this authorDamera Vakula, Damera Vakula Department of Electronics and Communication Engineering, National Institute of Technology Warangal, Warangal, Telangana, 506004 IndiaSearch for more papers by this author Ratikanta Sahoo, Corresponding Author Ratikanta Sahoo ratikanta.sahoo15@gmail.com orcid.org/0000-0001-5050-9292 Department of Electronics and Communication Engineering, National Institute of Technology Warangal, Warangal, Telangana, 506004 IndiaSearch for more papers by this authorDamera Vakula, Damera Vakula Department of Electronics and Communication Engineering, National Institute of Technology Warangal, Warangal, Telangana, 506004 IndiaSearch for more papers by this author First published: 26 July 2019 https://doi.org/10.1049/iet-map.2018.6008Citations: 7AboutSectionsPDF ToolsRequest permissionExport citationAdd to favoritesTrack citation ShareShare Give accessShare full text accessShare full-text accessPlease review our Terms and Conditions of Use and check box below to share full-text version of article.I have read and accept the Wiley Online Library Terms and Conditions of UseShareable LinkUse the link below to share a full-text version of this article with your friends and colleagues. Learn more.Copy URL Share a linkShare onFacebookTwitterLinkedInRedditWechat Abstract In this study, a novel metamaterial inspired compact cylindrical conformal dual-band antenna with a shunt fractal inductor and bottom patch is proposed. The conformal antenna consists of three meander lines type unit cells of composite right/left-handed metamaterials for modification and compactness. To improve the operating bandwidth in the first band i.e at 2.45 GHz, a shunt fractal inductor with large inductance is proposed. A similar patch with a slot along the bottom side is additionally used for better impedance matching. The second band at 3.5 GHz is obtained due to the loading of two meander line unit cells on the patch as parasitic elements. The proposed conformal antenna is placed on a 15 mm radius cylinder whose equivalent planar dimension is 0.48λg × 0.36λg (where λg is the guided wavelength). Asymmetrical coplanar waveguide feed is implemented. Measurement results showed that the proposed cylindrical conformal antenna achieves a wideband −10 dB fractional bandwidth of 44.5% and a gain of 1.26 dBi at the operating frequency of 2.45 GHz and 4.5% of bandwidth and a gain of 2.2 dBi at the 3.5 GHz. The radiation efficiency is 70 and 86% at the said frequencies respectively. There is good agreement between measured and simulation data. 1 Introduction Wireless communication plays a significant role in the day to day life. Different wireless communication technologies such as radar, satellite communication, cellular communication, global positioning system, Bluetooth, radio frequency identification, Wi-Fi and WiMAX are available for a variety of functions. Wi-Fi is a low-cost wireless technology, which serves as a router for communication to portable devices with an internet connection that allows the user to connect in close proximity. WiMAX is a broadcast communication system, which is used for wireless data transfer over long distances. It has potential application in military and commercial communications. Omnidirectional coverage antennas are required for indoor and outdoor wireless applications. Wi-Fi and WiMAX antennas are to be placed in a convenient location on the platform where the omnidirectional coverage is guaranteed without any obstruction of the signal. In any case, the platform for mounting the antenna is not always flat. In some instances, it is recommended to make the antenna conformal to the device to make it less visible to the human eye [[1]–[3]]. As space constraints on the device are always a limitation for antenna design, a compact dual-band structure is required. The microstrip patch antenna is the most suitable for microwave frequency band for modern wireless communications [[4]]. The miniaturisation of the patch antenna is the desired property and many efforts have been made for this purpose, such as using shorted pin [[5]–[7]], employing substrate with high-dielectric constant [[8]] and utilising loaded slots and capacitances [[9], [10]]. If the dielectric constant of the substrate is increased, the size of the antenna reduces but with an increase in quality factor and a decrease in operating bandwidth. Compact four-element split-ring resonator-loaded dual-band multiple-input multiple-output antenna is designed for the wireless local network (WLAN)/WiMAX/WiFi/4G-LTE and 5G applications in [[11]]. Compactness is achieved in an antenna by employing a modified meandered slot, a slotted Y-shaped monopole, and a trident-shaped feed strip in [[12]]. A dual-band circularly polarised (CP) planar monopole antenna is designed for WLAN/Wi-Fi/WiMAX/Bluetooth applications [[13]]. Research methods are developed to simultaneously increase the bandwidth and minimise the size of the patch antenna. This is a very challenging task but useful for many applications such as handset internal antennas, radiofrequency front-end antennas and antennas for millimetre waves. Moreover, miniaturisation will reduce system requirements [[14]–[16]]. Since the inception of metamaterials (MTMs), the demand for compact and wideband antenna has increased [[17]]. There is always a trade-off between bandwidth and size in the conventional antennas, which makes them unsuitable for platforms such as moving vehicles, missiles and hand held devices. MTMs give compactness by loading conventional antennas with single or numerous unit cells. Mode dispersion and Q factor are significantly influenced by MTM unit cells. This unique property gives rise to the miniaturisation of antennas. Recently, some studies on bandwidth enhancement and MTM antennas on the planar surface were published in the literature [[18]–[22]]. To enhance the bandwidth, Zhu and Eleftheriades implemented a transmission line MTM antenna of size operating at 3.3 GHz. A fractional bandwidth of 3.1% is attained by loading shunt spiral inductors but the design is complex due to the presence of vias [[18]]. Chi et al. designed a zero order resonance antenna of size operating at 2.16 GHz, using an interdigital capacitor as unit cell [[19]]. In this case, the meander lines type inductor is used, which has properties such as low inductance and self-resonance frequency (SRF). The fractional bandwidth obtained for the monopole antenna is of 15.1% but the actual bandwidth of the antenna is low. Ji et al. presented an antenna based on two planar mushroom structure unit cells [[20]]. The antenna which operates at 1.99 GHz has an overall size of . Fractional bandwidth of up to 20.3% is achieved by merging two resonance modes using an inductor and a capacitor. As the antenna is resonating at two adjacent modes, there is a possibility in the bandwidth reduction for a slight change in resonant modes. Moreover, the fractional bandwidth is comparatively lower and the size of the antenna is also large. Li et al. proposed a dual-band planar antenna loaded by a square patch with a slot on the bottom side. The antenna designed constitutes a composite right and left-handed (CRLH) MTM unit cell for WLAN/WiMAX applications [[21]]. The size of the antenna is and the bandwidths of the dual band are 22.8 and 10.8%, respectively. However, the size of the antenna is large with low bandwidth. A dual mode MTM antenna with vias was designed by Bala et al. by merging zeroth-order resonance (ZOR) mode and positive order resonance (POR) mode [[22]]. The bandwidth achieved is 39% for the size of , but the presence of vias makes the design complex. In this study, a new cylindrical conformal dual-band antenna for Wi-Fi and WiMAX applications is proposed. The novelty of the work is the design of a conformal dual-band antenna, which is devised with a 15 mm radius of the cylinder and this type of design is not reported in any literature published earlier. Finally, the MTM antennas published in the literature have narrow bandwidth and large size. Hence, in the proposed work, the concept of inductor loading is applied to achieve more bandwidth in the first band and by incorporating MTM because of which the size of the antenna is also reduced. The design of conformal dual-band antennas for two important bands such as Wi-Fi and WiMAX applications is a challenging task and is not available in published literature, hence an effort is made in the present study. The antenna bandwidth is increased by adding the first iteration circular fractal-shaped shunt inductor to the radiating patch. The input reflection coefficient is also improved due to the presence of a bottom patch with the slot. The presence of two meander lines as parasitic elements on the patch resonates the antenna in the second band at 3.5 GHz, which can be used for WiMAX applications. In the proposed work, meander line MTM unit cell is used as parasitic elements, which were not reported in earlier literature. Generally, the fabrication of conformal dual-band antenna is a challenging task. However, the conformal dual-band antenna is fabricated and there is good agreement between experimental and simulation results. The proposed cylindrical shaped conformal dual-band antenna radiates an omnidirectional pattern in H-plane and operates in the first frequency band of 1.69–2.66 GHz and also in the second frequency band of 3.42–3.58 GHz. 2 Design methodology of the antenna and parametric study 2.1 Design of conventional monopole conformal antenna A printed monopole antenna is a good candidate for wireless communication services because of its wide impedance bandwidth, omnidirectional radiation pattern, compact structure, and ease of fabrication. The geometry and parameters of the planar antenna are shown in Fig. 1. The antenna is fed by the coplanar waveguide (CPW) where the ground plane and radiating patch are present on the same side of the substrate for easy fabrication process as compared with a microstrip line feed, which requires ground plane on the opposite side of the patch. The ground plane and patch are placed on the same plane with the goal of via-free design. The shunt capacitance between the ground and top patch can be effectively balanced in CPW feed. Shunt capacitance ‘’ is formed between the radiating patch and CPW ground which is calculated as [[23]] (1) where is the relative permittivity, is the permittivity in free space, A is the area, and D is the gap between the patch and CPW ground. Fig. 1Open in figure viewerPowerPoint Structure of the conventional monopole planar antenna The antenna is designed at a frequency higher than Wi-Fi frequency (2.45 GHz) because by applying size reduction techniques, the resonant frequency is going to decrease to Wi-Fi frequency. RT/duroid® 5880 with a dielectric constant of 2.2, thickness of 0.787 mm and dielectric loss tangent of 0.0009 is selected as a substrate. The selected substrate is thermally stable and can be easily bent to make the conformal antenna. The antenna is designed and simulated using commercially available computer simulation technique (CST) software. A hexagonal shape antenna is initially selected as it gives a compact structure compared with other shapes. Modification of hexagonal shape is done to incorporate the MTM structures for achieving size reduction. So, the hexagonal shape is modified to octagonal. The dimensions of patches ‘’ and ‘’ are asymmetrical due to the optimised positions of the MTM unit cell. The dimension of ‘’ is decided by the size of the MTM unit cell to be introduced in the antenna design. The optimised dimensions of the planar antenna in Fig. 1 are shown in Table 1. The planar monopole antenna is transformed to the conformal antenna on a cylinder of radius 15 mm, which is indicated in Fig. 2. The input reflection coefficient varies with ‘’ distance between the lower edge of the radiating patch and upper edge of the ground plane. The input reflection coefficients of the conformal antenna with different values of ‘’ are displayed in Fig. 3. The optimised value of ‘’ as 2.5 mm is selected for the proposed antenna. Table 1. Optimised dimension of the proposed antenna Parameters Size, mm Parameters Size, mm L 40 2.5 W 30 2.4 14.9 11 4 13.45 7.52 g 0.35 10 — — Fig. 2Open in figure viewerPowerPoint Structure of the conventional monopole conformal antenna Fig. 3Open in figure viewerPowerPoint Input reflection coefficient of the conventional monopole conformal antenna with a different value of 2.2 Design of conformal antenna loaded with meander lines 2.2.1 CRLH transmission line (CRLH-TL) MTM MTMs are defined as artificial effective homogeneous electromagnetic structures with unusual properties not readily available in nature. An effectively homogenous structure is a structure whose average cell size is much smaller than the guided wavelength, i.e. the average cell size ‘p’, which must be smaller than a quarter of wavelength ‘’ . The parameters of the material such as permittivity and permeability depend on the unit cell structure and its periodic arrangement. The design of MTM antennas is based on the CRLH-TL unit cell shown in Fig. 4. It consists of a conventional host TL with series inductance , series capacitance , shunt inductance , and shunt capacitance . Fig. 4Open in figure viewerPowerPoint Equivalent circuit model of fundamental MTM unit cell The analytical dispersion equation of the CRLH-TL can be obtained by applying Bloch and Floquet theory to the unit cell as follows [[17]]: (2) where is the propagation constant for Bolch waves, and d is the length of the unit cell. The resonance of the CRLH-TL for resonance modes n can be obtained by the following condition: (3) where N and l are the number of the unit cell and the total length of the resonator, respectively. 2.2.2 Design methodology In the proposed work, the MTM concept is applied for compactness, where the size of the antenna can be miniaturised by modifying the patch structure with MTM unit cells. Although MTMs are periodic structures, in the conformal antenna shown in Fig. 5, only three meander lines type unit cells are used as in [[16], [19]]. Meander lines give a compact structure with an increase in current through the zigzag path of the lines. The parameters of the meander line unit cell are the width of the line, the gap between lines and number of fingers. The width of the lines controls the current flow, which in turn governs the inductance of the path. By increasing the number of fingers, the effective electrical length increases with the decrease of the resonant frequency. More than one unit cell can be incorporated to reduce the size of the antenna. Fig. 5Open in figure viewerPowerPoint Structure of the monopole conformal antenna loaded with meander lines In the present work, the meander line unit cell with a width of 0.2 mm, spacing gap of 0.2 mm and finger length of 5 mm is selected. The finger length (5 mm) is equivalent to 0.06 at 2.45 GHz and satisfies the condition for the average cell size, which must be smaller than a quarter of a wavelength, i.e. . In the equivalent circuit model indicated in Fig. 4, it can be interpreted that series inductance and series capacitance are decided by the meander lines. In this model, the series capacitance is the interdigital capacitance due to meander lines and the series inductance is due to the current flow on meander lines. The extracted values for parameters and in this model are given below [[23]] (4) (5) (6) where is the relative permittivity, l is the length of the finger, N is number of fingers, h is the height of the substrate, and W is the width of one finger (7) where is a number of turns, is the length of a lead, is the height of the meander, is the width of the meander, and is the width of the printed strip. To evenly distribute the meander line along with the patch structure, three unit cells are considered. To observe the effect of a number of fingers in the meander lines, the simulated input reflection coefficients for the proposed conformal antenna are shown in Fig. 6. It is found that the resonant frequency of the proposed antenna is decreasing from 3.4 to 2.76 GHz while increasing the number of fingers from 0 to 5 in the meander lines. It indicates that the antenna in Fig. 2 which is operating at 3.4 GHz before introducing meander lines is operating at 2.76 GHz after implementing meander lines, as shown in Fig. 5. Initially, the number of fingers is considered as four and on further increasing the number of fingers beyond four, it is observed that gain is decreased, as illustrated in Fig. 7. Fig. 6Open in figure viewerPowerPoint Simulated input reflection coefficient of the conformal antenna with a different number of fingers m Fig. 7Open in figure viewerPowerPoint Simulated gain of the conformal antenna with a different number of fingers m The dispersion characteristics of the proposed CRLH-TL unit cell obtained by using simulated S-parameters are shown in Fig. 8. The dispersion curve relates frequency and phase constant, where the curve depends on the inductance and capacitance values of CRLH-TL unit cells. It shows that there is a smooth transition from the right-handed pass-band and left-handed pass-band at ZOR of 2.45 GHz. From the dispersion characteristics, the proposed antenna can be realized as the ZOR antenna. The MTM-ZOR antenna can be realised using the CRLH transmission structure. Fig. 8Open in figure viewerPowerPoint Dispersion diagram of the proposed unit cell The Q factor and fractional bandwidth of the ZOR circuit are given in [[14]] (8) (9) Compared to conventional resonant antennas, ZOR antennas are known to have narrow bandwidth issue due to small and large . The antenna structure follows the open ended boundary condition. The ZOR frequency is controlled by distributed shunt elements such as and of the transmission line. With a specific end goal to enhance the bandwidth of the structure, material with low permittivity and high thickness is utilised. However, this reduces the design freedom and causes fabrication difficulties. An antenna with an increase in is suggested in the proposed work by adding inductance which results in enhanced bandwidth without diminishing the efficiency. 2.3 Design of conformal antenna loaded with the fractal inductor The antenna can be made compact by reducing the resonant frequency by lumped impedance loading. This is attempted by adding an additional inductor to the patch, which is not reported in the published literature. The inductor can be considered as strip inductor, circular geometry or meander line. Circular geometry has the best electrical performance, whereas other geometries such as meander line inductors are seldom used because of low inductance and SRF. The inductance of circular-shaped inductor can be increased by increasing the mean radius of the coil. A fractal inductor is proposed as shown in Fig. 9 when the mean radius of the coil is large which results in a large area of fractal shape when compared with inductor without fractal structure. Circular geometry can be modified as a fractal structure by the repetition of the circular boundary. In the proposed work, circular boundary and the first iteration of the circular boundary, which is named as a fractal inductor, are considered. Also, the proposed fractal structure can be manufactured easily. The presence of a fractal inductor is adding shunt inductance in the equivalent circuit model as given in Fig. 4 and can be calculated as [[23]] (10) (11) where a is the mean radius of a circular loop, W is the width of the loop, t is the thickness of the line, and h is the height of the substrate. Fig. 9Open in figure viewerPowerPoint Structures of the planar equivalent of the proposed asymmetric coplanar waveguide (ACPW) conformal antenna loaded with meander lines and fractal single turn inductor For the proposed antenna, equivalent circuit parameters are calculated using (1), (4), (7) and (10), which are obtained as , , and as shown in Fig. 4. The variation of simulated input reflection coefficients with frequency for the proposed conformal antenna with different shapes of the inductor are shown in Fig. 10 for comparison. The resonant frequency of an antenna with an inductor is decreased when compared with that without inductor. The antenna with a strip inductor and circular inductor has similar input reflection coefficient variation. However, first iteration fractal inductor has better input reflection coefficient at the resonant frequency with an increase in bandwidth. The fractional bandwidth obtained for the antenna without the inductor, strip inductor, single turn inductor, and the fractal inductor is 12, 43, 45, and 51.3%, respectively. The change in resonant frequency is due to the change in the electrical length of the current path. The fractional bandwidth of the proposed cylindrical conformal antenna with a fractal shaped inductor is high when compared to the strip and single turn inductor. Fig. 10Open in figure viewerPowerPoint Simulated input reflection coefficient of the proposed conformal antenna with different inductor shapes The location of the inductor on the patch also decides the resonant frequency. To observe this effect, the inductor is located at different heights and the simulated input reflection coefficients are plotted in Fig. 11. It is illustrated that the resonant frequency can be reduced by decreasing the height. When the height is 22.3 and 24.3 mm, the input reflection coefficient characteristics are similar, but at 24.3 mm the input reflection coefficient is the minimum; hence 24.3 mm location is selected as the optimised position. Fig. 11Open in figure viewerPowerPoint Simulated input reflection coefficient of the proposed conformal antenna with different position of the fractal inductor The simulated gain versus frequency of the conformal antenna without meander lines and fractal inductor, with meander lines only and the presence of both meander lines and fractal inductor, is compared in Fig. 12. It can be investigated that the gain of the proposed antenna is reduced due to compactness by introducing meander lines and fractal inductor in the antenna. Fig. 12Open in figure viewerPowerPoint Simulated gain versus frequency of the conformal antenna with and without meander lines and fractal inductor 2.4 Design of conformal antenna loaded with a slotted bottom patch To match the impedance, a novel bottom patch is proposed in Fig. 13. The bottom patch can be placed either on the lower side ‘’ or on the upper side ‘’. The input reflection coefficients at two locations on the bottom patch, with and without slot, are shown in Fig. 14. It can be observed that the bottom patch below with slot has less input reflection coefficient at the resonant frequency, hence it is considered in the proposed design. Fig. 13Open in figure viewerPowerPoint Bottom view of the planar equivalent of the proposed conformal antenna Fig. 14Open in figure viewerPowerPoint Input reflection coefficient of the proposed conformal antenna with effects of patch 2 The proposed antenna which is conformal on a cylindrical device of radius 15 mm is shown in Fig. 15. The input reflection coefficients of the conformal antenna on different radii of curvature are shown in Fig. 16. The small changes in the resonant frequency of the proposed cylindrical conformal antenna from the respective planar antenna are because of bending into conformal, resulting in a change in the effective resonant length. The resonant frequency decreases as the radius of curvature decreases, which is shown in Fig. 16. In the present study, the variations of the resonant frequency of the conformal antenna on cylinders of different radii are also established. The optimised dimensions of the planar version of the proposed conformal antenna are shown in Table 2. Fig. 15Open in figure viewerPowerPoint Proposed cylindrical conformal antenna with a 15 mm radius of curvature Fig. 16Open in figure viewerPowerPoint Comparison between simulated input reflection coefficient of the planar and conformal antenna with 5, 10 and 15 mm radius of curvature Table 2. Optimised dimension of the proposed conformal antenna Parameters Size, mm Parameters Size, mm L 40 2.5 w 30 9 14.9 4 4 16.2 7.52 S 0.2 10 1 0.5 0.8 2.4 0.6 11 0.4 13.45 D 1.6 g 0.35 22.3 2.5 Design of conformal dual-band antenna loaded with meander lines as parasitic elements on patch The dual-band characteristics of the proposed antenna are obtained by incorporating two meander lines unit cells as parasitic elements. A novel concept of meander lines as parasitic elements is placed in the antenna design for obtaining additional resonant frequency band. The parasitic elements are placed on the right and left most part of the top patch as shown in Fig. 17. The antenna is resonating at 2.45 GHz without parasitic elements as shown in Fig. 14. The proposed cylindrical conformal antenna is operating in the second band at 3.5 GHz along with the first band at 2.45 GHz as shown in Fig. 18 by loading meander lines as parasitic elements on patch. Fig. 17Open in figure viewerPowerPoint Proposed cylindrical conformal dual-band antenna with a 15 mm radius of curvature Fig. 18Open in figure viewerPowerPoint Input reflection coefficient of the proposed conformal dual-band antenna 3 Results and discussion The prototype of the antenna is fabricated using LPKF milling machine S100 on a flat substrate and it is rolled up on a cylindrical shape foam of radius 15 mm as shown in Fig. 19 and 20. The foam material is used as a mechanical support and has negligible radiation effect on the antenna. Fig. 19Open in figure viewerPowerPoint Photographs of fabricated antennas (a) Top view of the proposed flat antenna, (b) Bottom view of the proposed flat antenna, (c) Top view of a proposed cylindrical conformal antenna with a 15 mm radius of curvature Fig. 20Open in figure viewerPowerPoint Photographs of dual-band fabricated antennas (a) Top view of the dual-band planar antenna, (b) Bottom view of the dual-band planar antenna, (c) Top view of a proposed cylindrical conformal dual-band antenna with a 15 mm radius of curvature Both measured and simulated input reflection coefficients of the proposed cylindrical conformal antenna with and without parasitic elements of a 15 mm radius of curvature are illustrated in Fig. 21. The simulated input reflection coefficient is obtained using CST STUDIO SUITE 2016 by considering discrete port type with impedance. A four-hole round shaped pin with female type subminiature version A (SMA) connector, which is working in the range DC to 18 GHz is connected to the fabricated antenna. The antennas are measured utilising an HP vector network analyser within the frequency range of 130 MHz to 13 GHz. The wideband response of the conformal antenna without parasitic elements is achieved by properly choosing the physical dimensions of unit cells and fractal inductor. There is good agreement between the simulation and measurement results. The −10 dB input reflection coefficient bandwidth is around 1.17 GHz (1.75–2.92) GHz with a fractional bandwidth of 50.1%. The dual-band characteristic of the conformal antenna is achieved due to the presence of meander lines as parasitic elements. The −10 dB input reflection coefficient bandwidth of the first band is around 970 MHz (1.69–2.66) GHz with a fractional bandwidth of 44.5% and for the second band it is around 160 MHz (3.42–3.58) GHz with a fractional bandwidth of 4.5%. Fig. 21Open in figure viewerPowerPoint Input reflection coefficient of the proposed conformal antenna with and without parasitic elements The radiation patterns at 2.45 and 3.5 GHz are described in Figs. 22 and 23, respectively, for both H-plane and E-plane. The two-dimensional patterns of the antenna are obtained in an anechoic chamber. To measure the radiation pattern, the standard horn antenna is used at the transmitter and the conformal antenna is connected to the receiver. The conformal antenna radiation pattern is measured in receiving mode. A figure of eight shape radiation pattern is obtained in E-plane and omnidirectional pattern in H-plane at the 2.45 as well as 3.5 GHz, respectively. The surface current distribution at 2.45 and 3.5 GHz is shown in Figs. 24 and 25, respectively. It shows how the meander lines on the patch and parasitic meander lines contribute to radiation as the surface current is densely distributed in those areas. In addition, the fractal inductor is also contributing to the radiation part. The surface current distribution is also plotted in Fig. 25 at a frequency of 3.5 GHz for the same antenna to support this discussion. The phase variation of surface current on meander lines is also shown in Figs. 24 and 25. The gain of the antenna at 2.45 and 3.5 GHz is shown in Fig. 26 as 1.26 and 2.2 dBi, respectively. Fig. 22Open in figure viewerPowerPoint Radiation pattern at 2.45 GHz (a) E-plane pattern, (b) H-plane pattern Fig. 23Open in figure viewerPowerPoint Radiation pattern at 3.5 GHz (a) E-plane pattern, (b) H-plane pattern Fig. 24Open in figure viewerPowerPoint Surface current distribution of proposed conformal antenna at 2.45 GHz Fig. 25Open in figure viewerPowerPoint Surface current distribution of proposed conformal antenna at 3.5 GHz Fig. 26Open in figure viewerPowerPoint Gain of the proposed conformal antenna with and without parasitic elements The simulated and measured results of the radiation efficiency for the proposed conformal dual-band antenna are shown in Fig. 27. The radiation efficiency is calculated with the antenna gain (G) and directivity (D), , which is approximately around 75 and 90% at 2.45 and 3.5 GHz, respectively. The measured radiation efficiency is around 70 and 86% at 2.45 and 3.5 GHz, respectively, indicating that low-radiation losses are achieved by the antenna. In practical cases, some of the power fed to the antenna terminals is always lost. Moreover, energy is lost in the dielectric material of the antenna and creates unintended heat. Together, these losses lead to situations where the antenna radiated efficiency in actual operation, which is always less than the ideal simulation results. Fig. 27Open in figure viewerPowerPoint Radiation efficiency of the proposed conformal antenna with and without parasitic elements The overall performance of the proposed conformal dual-band antenna is compared with that of previously reported planar antennas [[11]–[13],[18]–[22]] in Table 3. The proposed antenna is realised without vias and it also provides competitively wide bandwidth in the first band. Moreover, the size is also more compact than for antennas published in [[11], [13], [18], [20], [21]]. Table 3. Comparison of the proposed cylindrical conformal dual-band antenna with planar antenna existing in the literature Antenna Size, mm2 Frequency, GHz −10 dB fractional bandwidth, % Gain, dBi Radiation efficiency, % Polarisation proposed work 40 × 30 2.45, 3.5 44.5, 4.5 1.26, 2.2 70, 86 linearly polarised (LP) [[11]] 40 × 40 2.93, 5.68 35.21, 6.86 4 89.55, 83.48 LP [[12]] 30 × 20 2.975 41.8 2 — CP [[13]] 42.5 × 48 2.38–2.75, 4.05–6.38 14.4, 44.67 0.88 to 1.4, 3.9 to 4.5 — CP [[18]] 50 × 50 3.30 3.1 0.79 65.8 LP [[19]] 32.2 × 20 2.16 15.1 1.62 72 LP [[20]] 40 × 100 1.99 20.3 1.05 to 3.35 41.5 to 66.6 LP [[21]] 70 × 44 2.5, 5.8 22.8, 10.8 1.5, 4.4 > 94, > 92.1 dual polarised [[22]] 39 × 25 2.5 (ZOR) and 3.32 (POR) 39 1.32 75 (ZOR), 91 (POR) LP 4 Conclusion A compact MTM-based cylindrical conformal dual-band antenna is designed, fabricated and validated experimentally. 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