Wideband bandstop filters based on wideband 180° phase shifters
2020; Institution of Engineering and Technology; Volume: 14; Issue: 13 Linguagem: Inglês
10.1049/iet-map.2020.0181
ISSN1751-8733
Autores Tópico(s)Antenna Design and Analysis
ResumoIET Microwaves, Antennas & PropagationVolume 14, Issue 13 p. 1662-1670 Research ArticleFree Access Wideband bandstop filters based on wideband 180° phase shifters Lei-Lei Qiu, orcid.org/0000-0002-3480-9292 Department of Electrical and Computer Engineering, Faculty of Science and Technology, University of Macau, Avenida da Universidade, Taipa, Macau SAR, People's Republic of ChinaSearch for more papers by this authorLei Zhu, Corresponding Author leizhu@um.edu.mo Department of Electrical and Computer Engineering, Faculty of Science and Technology, University of Macau, Avenida da Universidade, Taipa, Macau SAR, People's Republic of ChinaSearch for more papers by this author Lei-Lei Qiu, orcid.org/0000-0002-3480-9292 Department of Electrical and Computer Engineering, Faculty of Science and Technology, University of Macau, Avenida da Universidade, Taipa, Macau SAR, People's Republic of ChinaSearch for more papers by this authorLei Zhu, Corresponding Author leizhu@um.edu.mo Department of Electrical and Computer Engineering, Faculty of Science and Technology, University of Macau, Avenida da Universidade, Taipa, Macau SAR, People's Republic of ChinaSearch for more papers by this author First published: 24 September 2020 https://doi.org/10.1049/iet-map.2020.0181AboutSectionsPDF ToolsRequest permissionExport citationAdd to favoritesTrack citation ShareShare Give accessShare full text accessShare full-text accessPlease review our Terms and Conditions of Use and check box below to share full-text version of article.I have read and accept the Wiley Online Library Terms and Conditions of UseShareable LinkUse the link below to share a full-text version of this article with your friends and colleagues. Learn more.Copy URL Share a linkShare onEmailFacebookTwitterLinked InRedditWechat Abstract In this study, a class of wideband bandstop filters is proposed based on wideband 180° phase shifters. Different from traditional signal–interference-based bandstop filters, they make use of a 180° phase shifting in a wide operating band in principle. After numerical analysis of the phase responses of the two coupling-line paths, the optimal solutions for the largest 180° phase-shifting bandwidth or minimum impedance ratio can be obtained. Then, the wideband bandstop filter based on these two unique coupling lines is intensively analysed to reveal that three transmission zeros are generated in the stopband as the two paths have a 180° phase difference, thus exhibiting its wideband property. Moreover, by enlarging the impedance of the two paths, good out-of-band characteristics of the bandstop filter can be attained, while in-band performances are almost unchanged. Next, two types of bandstop filters with extended bandwidth are proposed by cascading an extra delay line for both paths, and they hold similar filtering characteristics as the aforementioned one. Finally, the design process and demonstration are provided. The measured results of the three fabricated filters show that Type-I, Type-II, and Type-III filters can satisfactorily obtain a 3 dB bandwidth of 108, 122, and 130% with 25, 14.7, and 11.6 dB rejection level, respectively. 1 Introduction Bandstop filters with wide and even ultra-wide bandwidth, high selectivity, and high rejection level are highly desired in many communication modules, such as oscillators and mixers, to remove harmonics and other spurious signals. To do so, one method is to use multimode resonators, such as stub-loaded resonators [[1], [2]]. However, this method is frequently used to design multiband bandstop filters. In order to realise a single stopband, two orthogonal eigenmodes of the dual-mode patch resonator were utilised [[3]]. Besides, bandstop filters using triple mode in a cubic dielectric resonator [[4]] and ceramic cavity resonator [[5]] were also proposed. However, this method on the multimode resonator is more suitable for a bandstop filter with a narrow band. To achieve a wideband stopband, the commonly used method is established through Kuroda's identity transformation from a low-pass topology response. This method can be implemented in the shunted/cascaded transmission-line type and spur-line type [[6]]. For the first type, it is easy to achieve a wide stopband by increasing the filter order. For the spur-line type, it has the inherent characteristic of compact and can be embedded between two adjacent shunted open stubs to introduce an extra attenuation pole. However, these methods are all based on lumped-element filters having maximally flat or Chebyshev responses. To achieve a flat stopband, elliptic-function [[7]] and pseudo-elliptic-function [[8]-[11]] bandstop filters were thus developed. As their common mechanism, the traditional shunted quarter-wavelength uniform-impedance branches are replaced with different lengths or different forms (such as stepped impedance) of branches, thereby generating more transmission zeros or attenuation poles at the real frequencies. Even though this method can achieve ultra-wide stopband, miniaturisation and steep selectivity, it unfortunately suffers from an intrinsic problem in the practical implementation of the connection line with high characteristic impedance. The third method is based on the coupling line structure. By using the cross-coupling of the branch line in the input and output ports, a quasi-elliptic function response with 80% fractional bandwidth (FBW) was realised [[12]]. In [[13]], based on a single quarter-wavelength resonator and one-section anti-coupled line, an extra-wide stopband with 120% 3 dB FBW and three attenuation poles was obtained. In [[14]], two lumped/distributed capacitors function as cross-coupling elements were adopted to acquire a 20 dB FBW of 110% with four attenuation poles. To generate more in-band poles, two-section anti-coupling lines [[15]] or multiple quarter-wavelength resonators [[16]] were studied for bandstop filters with enhanced selectivity and stopband rejection. Also, the hybrid structures of grounded resistors, branch line, and coupling lines were introduced to obtain wideband absorptive response [[17], [18]]. To acquire a wide stopband with high selectivity, the signal–interference structure is a very effective method, which is often applied to achieve highly selective passband [[19], [20]] or stopband [[21]-[28]]. To realise this kind of bandstop filters, the two transmission paths with a phase difference of 180° are connected in parallel at the input and output ports, such that the two transmission signals are eliminated so as to achieve a wideband bandstop characteristic. As shown in Fig. 1a, the phases of the two paths are indicated as φr and φm, respectively. The traditional signal–interference type considers a phase shift of 180° only at the centre frequency, that is φr(f) − φm(f) = 180° @ f0, which leads to a limited stopband bandwidth. In [[21], [22]], the two paths are composed of two transmission lines that differ by 180° in electrical length. Since the phase-shift condition of two transmission lines is only established at a single frequency f0, the achievable impedance bandwidth is limited up to 65%. In [[23]], the two paths both adopted small-sized coupling lines, and five attenuation poles were generated in the stopband of over 70% FBW. To further widen the bandwidth, one transmission-line path of the two-line signal–interference structure was replaced by a T-shaped branch [[24]] or coupling line [[25]], thus achieving about 100% FBW with 20 dB attenuation. Besides, a hybrid structure based on a coupling line path and a branch line path was presented [[26]-[28]], and its FBW is further expanded to 141.7%. Although the above signal–interference structure can realise a wide or even ultra-wide stopband, the design procedure is mainly executed based on the numerical analysis of the amplitude, and the phase analysis of its two paths has been rarely reported. Fig. 1Open in figure viewerPowerPoint Different type of signal–interference-based bandstop filter (a) Traditional one via a 180° phase shift at a single frequency, (b) Proposed one via a wideband 180° phase shift In this paper, wideband bandstop filters based on 180° wideband phase shifters are proposed and analysed. Its basic diagram is depicted in Fig. 1b. Herein, the two paths have a wideband phase shift of 180° ± δ in the fL–fU range, so as to achieve a wideband signal cancellation effect. Compared with the conventional one, this work focuses itself on the analysis and design of these wideband bandstop filters from the phase perspective for the first time. The remainder of this paper is organised as follows: in Section 2, two types of coupling lines are proposed, their phase characteristics are analysed, and the optimal solutions for wideband or minimum impedance ratio are given. In Section 3, a kind of signal–interference bandstop filter based on two types of coupled lines is proposed. The phase characteristics of these two coupling lines and the amplitude characteristics of the bandstop filters are analysed. After the effects of the impedances on the selectivity of the bandstop filters are investigated, two types of improved bandstop filters are proposed and designed in Section 4. In Section 5, the design process and design graph are demonstrated for a quick design. In Sections 6 and 7, experimental results and conclusions are provided. 2 Analysis of phase-shifting network Fig. 2 shows the schematics of the two types of transmission-line phase-shifting networks. The first type, as provided in Fig. 2a, is a two-section cascaded coupling line with the same image impedances Zm while different impedance ratios ρm1 and ρm2. Fig. 2b is a C-section coupling line which is short circuited at one end, and its image impedance and impedance ratio are Zr and ρr, respectively. These two coupling lines, also called Type-F and Type-C coupling lines in [[29]] can properly control the phase slope by the impedance ratio and are often used in the design of phase shifters. In [[30]], wideband phase shifters were proposed by utilising the coupling line working in the first and second phase periods. However, the above analysis is based on a single case, and there is no detailed analysis on how to obtain the optimal bandwidth and minimum impedance ratio of the phase shifter using these two types of coupled lines. In this section, the optimal solution of the 180° phase shifters will be described in detail. Fig. 2Open in figure viewerPowerPoint Schematics of two general Schiffman coupled lines (a) Two-section coupled line, (b) C-section coupled line For the coupled line in Fig. 2a, the two sections of the coupling line have the same image impedance, so the insertion phase shift can be expressed as (1a) (1b) For the coupled line in Fig. 2b, i.e. a traditional Schiffman coupling line, its insertion phase shift is (2) Therefore, the phase difference or phase-shift value of the two types of coupling structures is ΔΦ(θ) = Φr(θ) – Φm(θ) (3) To generate a constant 180° phase shift, the insertion phase shift and phase-shift slope need to meet the requirements around the centre frequency as (4a) (4b) where ρm1, ρm2, and ρr are impedance ratios to be defined as (5a) (5b) (5c) Through (4b), all solutions can be obtained as demonstrated in the 3D plot in Fig. 3a. By making use of the cross-section view in the ρm1ρr plane, further analysis can be obtained as shown in Fig. 3b. It can be seen herein that when ρm2 is fixed, the solution composed of ρm1 and ρr is an inversely proportional curve, that is the larger ρm1 is, the smaller ρr is, and vice versa. When ρm2 increases, the curve of the solution composed of ρm1 and ρr moves to the upper right corner. Apparently, in the case of minimum impedance ratio of ρmax = max {ρm2, ρm1, ρr}, ρm2 = 1 and ρm1 = ρr, i.e. case B, becomes an optimal solution. It should be noted here that the minimum of ρmax means that the maximum impedance ratio among ρm2, ρm1, and ρr is the smallest, thus the three coupling lines are most likely physically realised. Fig. 3Open in figure viewerPowerPoint Calculated solutions of a 180° phase shift for the two types of coupled lines (a) 3D presentation of all solutions, (b) Cross-section view in ρm1ρr plane, (c) Phase shift response of the two types of coupled lines under different cases A–F Additionally, when ρmax is less than or equal to a constant value, for example 3, all the solutions of ρm2, ρm1, and ρr are in the shaded area enclosed by the curves A–F, where the corresponding parameters under A–F cases are tabulated in Table 1. Fig. 3c plots the phase-shift curves of the two types of coupled lines under different parameters. It can be seen that the phase shift at the central frequency is fixed at 180° in different cases. When ρm2 keeps constant, and the parameters ρm1 and ρr vary with the trend of A–B–C or D–E–F, the phase-shift bandwidth gradually falls down. As the parameters vary from A to D, the phase-shift bandwidth tends to gradually decrease. It can be concluded that the phase-shifting bandwidth is the largest in case A, where ρm2 = 1 and ρr = 1. Table 1. Parameters for cases A–F Cases ρm2 ρm1 ρr A 1 2.6 1 B 1 1.755 1.755 C 1 1.213 3 D 3 3 1.333 E 3 2.1745 2.1745 F 3 1.766 3 From the above analysis on the two types of coupled lines, when ρm2 = 1 and ρm1 = ρr, ρmax = max{ρm2, ρm1, ρr} is the minimum; when ρm2 = 1 and ρr = 1, the achievable phase-shifting bandwidth is the widest. In the following, the analysis will be carried out based on ρm2 = 1, where the phase-shifting bandwidth is adjusted by ρm1 (also defined as ρm) and ρr. 3 Wideband bandstop filters (Type-I) Based on the above analysis, the two types of coupled lines in Fig. 2 are in parallel shunted to form a signal–interference-type bandstop filter, as depicted in Fig. 4a. Fig. 4b shows the even- and odd-mode equivalent circuits. In the even-mode case, the input impedance Zine of the equivalent circuit is Zine = Zine1||Zine2, where (6a) (6b) Similarly, the input impedance Zino of the odd-mode equivalent circuit is Zino = Zino1 || Zino2, where (7a) (7b) where Zem/Zom denotes the even-/odd-mode input impedance of the uncoupled-line section of the Type-F coupled line. Fig. 4Open in figure viewerPowerPoint Schematic of the proposed signal–interference-based bandstop filters (Type-I) (a) Circuit model, (b) Even- and odd-mode half bisection Therefore, the S-parameter and transmission zero equation can be derived as (8) (9) Accordingly, the transmission zero can be calculated as (10a) (10b) (10c) where Through numerical methods, it can be known that three of the transmission zeros have the same solutions as the phase function (3). Therefore, these three transmission zeros are generated as a 180° phase shift of the two paths. Also, the other two transmission zeros are generated by the resonator formed by the two transmission paths. Also when Zm = Zr, these five transmission zeros are a function of the impedance ratios ρm and ρr, regardless of the specific impedance value Zm and Zr. Fig. 5a shows the insertion phase-shift curves of the two paths and the amplitude responses of the bandstop filters in three different cases. The parameters of these three cases are listed in Table 2. As can be seen in each case, the phase-shift curve indeed has three frequency points of 180°, corresponding to three transmission zeros or attenuation poles of the bandstop filter. When the case is changed from A1 to B1 then C1, ρm decreases and ρr increases gradually, the insertion phase-shift bandwidth and phase deviation of the two paths gradually decrease, which is the same as the analysis in Section 2. Accordingly, the bandwidth of the bandstop filter decreases dramatically, while the out-of-band selectivity becomes better. Table 2. Parameters for cases A1–F1(ρm2 = 1) Cases ρm ( = ρm1) ρr Z = Zm = Zr, Ω A1 3.237 1 100 B1 1.905 1.905 100 C1 1 4.493 100 D1 1.905 1.905 80 E1 1.905 1.905 100 F1 1.905 1.905 120 Fig. 5Open in figure viewerPowerPoint Phase responses of the two paths of the bandstop filter and magnitude responses of the bandstop filter (Type-I) (a) Cases A1–C1, (b) Cases D1–F1 Fig. 5b plots the insertion phase-shift curves and the amplitude responses of the bandstop filter when the impedance ratios ρm and ρr are constant and Zm = Zr varies. When the impedance value increases, the impedance values Zm and Zr of the two paths are always equal to each other, so the phase-shift curve remains unchanged. On the one hand, the positions of these transmission zeros of the bandstop filter get no change. In this context, the stopband has less fluctuation but remains almost unchanged stopband bandwidth. On the other hand, the impedance value may affect the matching property in the passband, so the out-of-band selectivity of the bandstop filter becomes better with the increase of the impedance. These characteristics are of great significance to the design of highly selective bandstop filters. 4 Bandstop filters with extended bandwidth (Type-II and Type-III) For the two types of coupled lines shown in Fig. 2, cascading transmission lines with the same characteristic impedance cannot change the phase difference between them but may affect the amplitude characteristics of the corresponding bandstop filter. In this section, this exhibited principle will be employed to design bandstop filters with extended bandwidth. As depicted in Figs. 6a and b, the input and output ports of Type-F (Type-C) coupled lines are, respectively, cascaded to the transmission lines with electrical length Kθ (Kθ) and impedances Zm (Zr). Since the coupled line and the transmission line are of the same impedance, good impedance matching is readily achieved. Therefore, the insertion phase-shift functions of these two structures are the same as (3). By utilising these two modified types of coupled lines as two parallel paths, a signal–interference-type bandstop filter can be realised, as depicted in Fig. 6c. Fig. 6d gives the even-mode and odd-mode equivalent circuits of the presented bandstop filter. Fig. 6Open in figure viewerPowerPoint Schematics of two general Schiffman coupled lines with cascaded delay lines (a) Two-section coupled line, (b) C-section coupled line, (c) Schematic of the proposed signal–interference-based bandstop filter with extended bandwidth (Type-II and Type-III), (d) Even- and odd-mode half bisection of the filter in (c) The even-mode input impedance of the bandstop filter is Zine = Zine1||Zine2, where (11a) (11b) (11c) (11d) The odd-mode input impedance of the bandstop filter is Zino = Zino1||Zino2, where (12a) (12b) (12c) (12d) Similarly to (8), the S-parameter and transmission zeros can be obtained, while different values K result in different types of bandstop filters which will be discussed as follows. Case 1: When K = 0.5, the bandstop filter is referred to as Type-II and its transmission zero equation is given as (13) As such, its transmission zeros can be deduced as (14a) (14b) (14c) where Case 2: When K = 1, the bandstop filter is referred to as Type-III, and its transmission zero equation is (15) where As can be known from (13) to (15), Type-II and Type-III filters can obtain five and seven transmission zeros in the desired stopband, respectively. Since the two paths of Type-I, Type-II, and Type-III bandstop filters have the same phase difference under the same parameters, three transmission zeros are the same, as can be verified by (10), (14), and (15). Except that these three transmission zeros are the same, the other transmission zeros are different. Fig. 7 shows the minimum and maximum transmission zeros of the three types of filters. It can be seen that when ρr = 1, the highest transmission zero frequency achieved by Type-II (Type-III) is higher than that of Type-I (Type-II). Due to the symmetrical property in frequency response, Type-II (Type-III) can achieve the lowest transmission zero less than that of Type-I (Type-II). This conclusion also holds when ρr = ρm. Consequently, compared to Type-I, the proposed Type-II and Type-III bandstop filters can indeed expand the bandwidth to a large extent. Fig. 7Open in figure viewerPowerPoint Graphical demonstration of the highest and lowest transmission zeros of the three types of filters Similarly, for Type-II filter, the bandwidth can be controlled by adjusting the values of ρm and ρr. Fig. 8a shows the insertion phase-shift curves of two paths and the amplitude responses of the bandstop filter in three different cases, and their parameters are given in Table 3. It can be seen that when the phase shift is equal to 180°, the bandstop filtering response has a corresponding transmission zero. Similarly to the same tendency analysed in Section 2, ρm1 decreases and ρr increases gradually when the parameters vary from A2–B2–C2. The insertion phase-shift bandwidth and phase deviation of the two paths thus decrease gradually. As a result, the bandwidth of the bandstop filter decreases gradually, while the out-of-band selectivity becomes better. Table 3. Parameters for cases A2–F2 (ρm2 = 1) Cases ρm ( = ρm1) ρr Z = Zm = Zr, Ω A2 4.6 1 100 B2 2.235 2.235 100 C2 1 6.01 100 D2 2.235 2.235 80 E2 2.235 2.235 100 F2 2.235 2.235 120 Fig. 8Open in figure viewerPowerPoint Phase responses of the two paths of the bandstop filter and magnitude responses of the bandstop filter (Type-II) (a) Cases A2–C2, (b) Cases D2–F2 To further improve the frequency selectivity of the stopband, the impedance values of the two paths can be properly adjusted. As demonstrated in Fig. 8b, when the impedance value increases, the phase difference responses of the two paths get no change, so the stopband has less fluctuation while keeping almost unchanged bandwidth. Additionally, the out-of-band selectivity of the bandstop filter is gradually improved. The comparison of the responses among the three types of bandstop filters is illustrated in Fig. 9. Type-III filter generates seven transmission zeros in the wide stopband with further widened bandwidth. For the general topology under the scenario with K ≥ 1, the analysis is similar to the above parts, and it will not be repeated here again. Fig. 9Open in figure viewerPowerPoint Phase responses of the two paths and magnitude responses of the bandstop filter under different types (Type-I: Zm = Zr = 130 Ω, and ρr = ρm = 1.879; Type-II: Zm = Zr = 122 Ω, and ρr = ρm = 2.175; Type-III: Zm = 121 Ω, Zr = 132.5 Ω, and ρr = ρm = 2.78) 5 Design approaches and curves For the proposed bandstop filters, the characteristic function responses are shown in Fig. 10. The design process of the bandstop filter is summarised as follows: (i) Specify the design parameters, such as bandwidth FBW. Fig. 10Open in figure viewerPowerPoint Characteristic functions of the three types of bandstop filters (ii) Determine θc, where θc is the lower cutoff frequency and FBW = (2−4θc/π) × 100%. (iii) Set Zm = 100 Ω, and obtain the initial values by numerical methods according to (16): (16) where θi is the local maximum points of |F|, i denotes 1, 2, and 3, and the characteristic function F is obtained by (17a) (17b) where RL is the rejection level in dB. Five (seven) unknown parameters ρm1, ρr, ɛ, θ1, and θ2 (ρm1, Zr, ρr, ɛ, θ1, θ2, and θ3) within five (seven) functions. (iv) Check whether the out-of-band selectivity meets the requirements. If not, increase the value of Zm and repeat the previous step to obtain the initial value until the out-of-band selectivity requirements are met. (v) Obtain the final physical size according to the mapping relationship. According to the above method, all the design parameters of the three types of filters, including bandwidth, suppression level, impedance, and impedance ratio are given in Figs. 11-13 as a guide for a quick and efficient design. Fig. 11Open in figure viewerPowerPoint Design parameters for the bandstop filter Type-I (a) FBW and rejection level, (b) ρr and Z Fig. 12Open in figure viewerPowerPoint Design parameters for the bandstop filter Type-II (a) FBW and rejection level, (b) ρr and Z Fig. 13Open in figure viewerPowerPoint Design parameters for the bandstop filter Type-III (a) FBW and rejection level, (b) ρr and Z 6 Results and discussion Based on the proposed topology, three bandstop filters (Type-I, Type-II, and Type-III) are designed, and their relative bandwidths are set as 77, 107, and 123%, respectively. The selected substrate has a thickness of 0.8 mm, a dielectric constant of 2.55, and a loss tangent of 0.0029. Its implemented structure is shown in Fig. 14, where the cutting portion in the coupled line is used to compensate for the unqual phase velocities of the odd- and even mode in the coupled microstrip line. The used electromagnetic simulation software is Keysight Advanced Design System (ADS), and the measured instrument is a vector network analyser R&S ZNB20. Fig. 14Open in figure viewerPowerPoint Physical layout and photograph of the filters (a) Physical layout, (b) Fabricated bandstop filters (Type-I, Type-II, and Type-III from right to left) For Type-I bandstop filter, the characteristic parameters are set as: FBW = 77%, f0 = 3 GHz, Zm = Zr = 130 Ω, and ρr = ρm = 1.879. The finally adjusted physical dimensions are shown in Table 4. Fig. 15a shows the simulated and measured results. It can be seen that five transmission zeros appear at 1.86, 2.45, 3.17, 3.73, and 4 GHz. The 3 dB (20 dB) bandwidth of the bandstop filter covers a frequency range of 1.44–4.67 GHz (1.73-4.22 GHz), that is FBW3 dB = 108% (FBW20 dB = 83%). The in-band rejection level is larger than 25 dB. The bandstop filter has good passband transmission characteristics, the return loss coefficient in the passband is better than 17.7 dB, and its maximum insertion loss in the lower passband within the DC–1.07 GHz range is 0.3 dB, and the maximum insertion loss of the upper passband in the range of 4.88–7.25 GHz is 1 dB. Table 4. Physical sizes of the proposed bandstop filters (unit:mm) Cases L1 L2 L3 L4 L5 L6 L7 Lm Lr W1 W2 W3 W4 W5 Wm Wr S1 S2 Type I 17.1 0 0 17.8 7.0 12.0 17.7 1.84 1.84 0.88 — — 0.3 0.28 0.62 0.62 0.29 0.29 Type II 17.2 9.1 9.1 17.9 7.7 14.2 17.8 1.84 1.84 0.72 0.31 0.31 0.26 0.31 0.44 0.44 0.21 0.21 Type III 17.5 18.3 17.9 17.1 6.31 14.4 18.75 2.74 2.74 0.45 0.28 0.36 0.28 0.36 0.3 0.19 0.13 0.13 Fig. 15Open in figure viewerPowerPoint Synthesised, simulated, and measured results of the proposed bandstop filters (a) Type-I, (b) Type-II, (c) Type-III For Type-II bandstop filter, the initial design parameters are: FBW = 107%, f0 = 3 GHz, Zm = Zr = 122 Ω, and ρr = ρm = 2.175. The final physical dimensions are included in Table 4. From the measured results in Fig. 15b, it can be seen that the maximum insertion loss in the lowpass band within the range of DC–0.98 GHz is 0.33 dB, and the maximum insertion loss of the upper passband is 1.1 dB in the range of 5.1–6.97 GHz. The stopband insertion loss and passband return loss are 14.7 and 13 dB, respectively. The 3 dB bandwidth of the bandstop filter is 1.23–4.89 GHz (FBW3dB = 122%), in which five transmission zeros can be observed intuitively. For Type-III bandstop filter, the initial design parameters are: FBW = 123%, f0 = 3 GHz, Zm = 121 Ω, Zr = 132.5 Ω, and ρr = ρm = 2.78. The final physical size and measured results are given in Table 4 and Fig. 15c, respectively. The bandstop filter has seven transmission zeros positioned at 1.23, 1.73, 2.37, 3.23, 3.83, 4.42, and 4.76 GHz. The 3 dB bandwidth frequency range of the bandstop filter is 1.06–4.97 GHz (FBW3dB = 130%). The insertion loss in the stopband and return loss in the passband are 11.6 and 11.3 dB, respectively. In addition, the maximum insertion loss at the lower passband up to 0.89 GHz is 0.39 dB, and the maximum insertion loss at the upper passband in the range of 5–6.9 GHz is 2.89 dB. There are slight deviations between simulation and measurement, which need to be pointed out herein. Firstly, for Type-II filter, there is a certain frequency shift in the upper skirt of the stopband, which may be due to the influence from the copper thickness, dielectric constant, and processing error. Secondly, the suppression level in stopband and return loss in passband increase as the bandwidth increases. This is mainly due to the insufficiently large external coupling. To address this problem, a conventional effective method is used to connect stubs in parallel at the input and output ports [[21]]. Thirdly, a spike appears in the upper passband around the second harmonic frequency, which has an impact on the insertion loss and group delay of the upper passband. It is primarily caused by the inconsistency of odd-mode and even-mode phase velocities in quasi-transverse electro-magnetic (TEM) microstrip coupling lines [[15]]. Although the compensation method is used, the design does not consider sufficient tolerance for processing error. Finally, Table 5 is tabulated for performance comparisons against those in other reported bandstop filters in state of the art. Compared with the published literature based on signal–interference, the proposed method is innovatively designed from the perspective of the wideband phase shifter, and a wider stopband is indeed realised. Overall, the proposed method is to extend the well-known signal–interference technique towards an alternative solution for the design of wideband bandstop filters. Table 5. Performance comparison with previous works Ref. Method TZs TPs ILmax in lower passband, dB ILmax in upper passband, dB 3 dB-FBW, % Attenuation in stopband, dB RL in passband, dB Sizea, λg2 [[8]] stub 3 4 0.24a 0.62a 138 21 14.7 — [[9]] stub 5 6 0.29a 2.3a 148 31 13.7 0.8 × 0.42 [[10]] stub 3 4 1 2 159 20 12.4 0.32 × 0.19 [[12]] stub + cross-coupling 3 6 0.75 2 64.7a 22.9 11.4 1.2 × 0.45 [[13]] coupled line + stub 3 4 0.4a 1a 120 20 18 0.12 × 0.51 [[14]] coupled line + cap 4 2 0.5 2 136a 20 18.3 0.45 × 0.061 [[15]] coupled line + stub 5 4 0.15 1.53 147a 34.4 17 0.738 × 0.054 [[17]] coupled line 5 8 1 1 70a 17 24 0.27 × 0.69 [[21]] signal–interference 3 4 0.8 1 50.6 41 10.4 0.116 × 0.163 [[23]] signal–interference 5 4 0.4 1.4 92 28.6 17.2 — [[24]] signal–interference 3 6 1 2 113 20 14.4 0.32 × 0.65 [[25]] signal–interference 4 4 1 2 126 19.6 10.1 0.35 × 0.31 [[27]] signal–interference 3 6 0.5a 2.5a 113 20 15 0.69 × 0.51 5 6 0.8a 2.5a 115 25 15 1.07 × 0.51 7 6 0.7a 3.7a 112.3 20 10 0.69 × 0.6 This work signal–interference + wideband phase shifter 5 4 0.3 1 108 25 17.7 0.2 × 0.61 5 4 0.33 1.1 122 14.7 13 0.2 × 0.72 7 4 0.39 2.89 130 11.6 11.3 0.3 × 0.85 TZ/TP: transmission zeros and poles. IL/RL: the insertion-/return-loss level. λg: the guided wavelength at the centre frequency f0 aEstimated values 7 Conclusion This paper proposes a class of wideband and ultra-wideband bandstop filters based on wideband 180° phase shifters. The filters utilise two parallel-connected Schiffman Type-C and Type-F coupled lines so as to construct its two distinct paths. Numerical analysis of the phase response of these two paths is conducted to obtain the optimal solutions for the largest 180° phase-shifting bandwidth or minimum impedance ratios. Compared with conventional signal–interference-based bandstop filters where a 180° phase shift only at a single frequency is considered, the two paths of the proposed one are expected to achieve a wideband 180° phase shift over a large frequency range. Furthermore, three types of filters with wideband and ultra-wideband stopband are presented and analysed. Three frequency points of 180° phase differences are matched well with the positions of three transmission zeros, furtherly revealing the wide stopband characteristics of the proposed bandstop filters. Moreover, their out-of-band property is improved by properly changing the path impedance under unchanged transmission zeros. These features are well suitable for the design and exploration of wideband high-selective bandstop filters. Finally, three types of bandstop filters are designed and fabricated for demonstration. 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