Low‐profile and wideband gain enhanced Fabry–Perot cavity antenna using gradient PRS and AMC
2020; Institution of Engineering and Technology; Volume: 14; Issue: 15 Linguagem: Inglês
10.1049/iet-map.2019.1129
ISSN1751-8733
AutoresZhi‐Peng Li, Lin Peng, Jing Ma, Bin Shi, Qixiang Zhao, Xing Jiang, Si‐min Li,
Tópico(s)Metamaterials and Metasurfaces Applications
ResumoIET Microwaves, Antennas & PropagationVolume 14, Issue 15 p. 1952-1959 Research ArticleFree Access Low-profile and wideband gain enhanced Fabry–Perot cavity antenna using gradient PRS and AMC Zhi-peng Li, Zhi-peng Li Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of ChinaSearch for more papers by this authorLin Peng, Corresponding Author Lin Peng penglin528@hotmail.com orcid.org/0000-0002-1255-3118 Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of China School of Physics, University of Electronic Science and Technology of China, Chengdu, 541004 People's Republic of ChinaSearch for more papers by this authorJing Ma, Jing Ma Science and Technology on special system simulation laboratory, Beijing Simulation Center, Beijing, 100854 People's Republic of ChinaSearch for more papers by this authorBin Shi, Bin Shi Science and Technology on special system simulation laboratory, Beijing Simulation Center, Beijing, 100854 People's Republic of ChinaSearch for more papers by this authorQi-xiang Zhao, Qi-xiang Zhao Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of ChinaSearch for more papers by this authorXing Jiang, Xing Jiang Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of ChinaSearch for more papers by this authorSi-min Li, Si-min Li Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of China School of Computer Science and Communication Engineering, Guangxi University of Science and Technology, Liuzhou, 545006 Guangxi, People's Republic of ChinaSearch for more papers by this author Zhi-peng Li, Zhi-peng Li Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of ChinaSearch for more papers by this authorLin Peng, Corresponding Author Lin Peng penglin528@hotmail.com orcid.org/0000-0002-1255-3118 Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of China School of Physics, University of Electronic Science and Technology of China, Chengdu, 541004 People's Republic of ChinaSearch for more papers by this authorJing Ma, Jing Ma Science and Technology on special system simulation laboratory, Beijing Simulation Center, Beijing, 100854 People's Republic of ChinaSearch for more papers by this authorBin Shi, Bin Shi Science and Technology on special system simulation laboratory, Beijing Simulation Center, Beijing, 100854 People's Republic of ChinaSearch for more papers by this authorQi-xiang Zhao, Qi-xiang Zhao Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of ChinaSearch for more papers by this authorXing Jiang, Xing Jiang Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of ChinaSearch for more papers by this authorSi-min Li, Si-min Li Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology, Guilin, 541004 Guangxi, People's Republic of China School of Computer Science and Communication Engineering, Guangxi University of Science and Technology, Liuzhou, 545006 Guangxi, People's Republic of ChinaSearch for more papers by this author First published: 22 October 2020 https://doi.org/10.1049/iet-map.2019.1129Citations: 3AboutSectionsPDF ToolsRequest permissionExport citationAdd to favoritesTrack citation ShareShare Give accessShare full text accessShare full-text accessPlease review our Terms and Conditions of Use and check box below to share full-text version of article.I have read and accept the Wiley Online Library Terms and Conditions of UseShareable LinkUse the link below to share a full-text version of this article with your friends and colleagues. Learn more.Copy URL Share a linkShare onFacebookTwitterLinked InRedditWechat Abstract Low-profile and wideband gain enhancement Fabry–Perot cavity antennas (FPCA) is proposed using gradient Partially Reflective Surface (PRS) and gradient Artificial Magnetic Conductor (AMC). A wideband source antenna is designed with a −10 dB impedance bandwidth of 58.1% (6.1–11.1 GHz). The PRS is constructed by a complementary frequency-Selective Surfaces (FSS) with a square hole and four small patches in a unit, which realizes phase increasing with frequency. The PRS and the AMC are gradient alone x-direction to achieve wideband phase compensation. While they are uniform in y-direction to realize better gain enhancement. Compared with the source antenna, the gain enhancement of the proposed FPCA is about 4–7 dBi. Moreover, the simulated and measured 3-dB gain bandwidths are 7.26–10.33 GHz (34.8%) and 6.8–10.8 GHz (46%), respectively. The gain enhanced bandwidth is about 52.3% (6.5–11.1 GHz). The impedance bandwidth is 6–11.1 GHz (59.65%). By using the gradient size AMC, the profile of the FPCA is reduced from 0.5λ0 to 0.2λ0, where λ0 is the wavelength of 8.5 GHz. The proposed FPCA has the characteristics of wide impedance bandwidth, wide 3-dB gain bandwidth, wide gain enhanced bandwidth and low-profile. 1 Introduction Compared with bulky horns or corporate-fed arrays, the Fabry–Perot cavity antenna (FPCA) has attracted significant attention due to its merits of high-gain performance, low profile, simple feeding network, fabrication simplicity and cost-saving potential [[1]]. Thus, the FPCA can be applied to numerous applications in modern communications, radar and sensor systems [[2]]. A conventional FPCA contains a partially reflective surface (PRS) placed at a half wavelength from the ground plane of the source antenna. Electromagnetic waves radiated from the source experience multiple reflections and transmissions within the cavity, which is formed by the ground plane and the top PRS [[3]]. As certain resonance conditions are satisfied, the waves transmitted through the PRS can be in phase [[4]]. Thus, significant gain enhancement can be achieved with this unique structure [[5]-[9]], which can be interpreted by ray theory [[10]]. Over the past few years, there are three major research interests on FPCA. The first one is reducing the profile of the FPCA [[11]-[13]]. To achieve this goal, the reflection phase of the ground plane should be tailored, such as 0°. The structure of in-phase reflection can be realised by loading artificial magnetic conductor (AMC), electromagnetic bandgap (EBG) etc. By using AMC with a 0° reflection phase, the overall FPCA profile can be reduced to ∼λ/4 [[11]]. In [[12]], by using a PRS–AMC superstrate (assembling) and filling the FPCA cavity with a dielectric substrate, the profile was reduced to λ/9. However, this method has a large loss. Although the phase engineering methods of the ground plane effectively reduce the profile of the FPCA, their gain bandwidths and impedance bandwidths are narrow. The second research interest is constructing circular polarised FPCA (CP-FPCA) [[14], [15]] or implementing multi-polarisation FPCA [[16]]. However, these designs still suffer from narrow bandwidths. The third research interest is conquering the defect of the narrow bandwidth of the traditional FPCA. As the FPCA resonance condition is usually realised in a narrow bandwidth, thereby, the impedance and gain bandwidths of the FPCAs are narrow [[4]]. Compared with the above researches, broadening gain bandwidth attracts many attentions. There are three main approaches to expand the gain bandwidth. One is to assemble multi-layer superstrates, which make the PRS reflection phase increasing with frequency [[2], [17]]. The experimental results in [[2]] show that the 3-dB gain bandwidth ranges from 13.5 to 15.7 GHz (15%) and its profile is 1.57λ. In [[17]] the 3-dB gain bandwidth ranges from 8.5 to 11.2 GHz (27.4%), and its profile is 1.23λ. The second method employs a single layer frequency-selective surface (FSS) structure [[5], [18]], such as complementary FSS structure. The complementary FSS in [[5]] realises reflection phase increasing with frequency, while the 3-dB gain bandwidth of the FPCA ranges from 8.6 to 11.4 GHz (28%). In [[18]], the 3-dB gain bandwidth ranges from 6.4 to 7.7 GHz (19.2%). However, the profile of these FPCAs are high and they are about 0.5λ. The third method constructs special structure, such as a wedge ground plane, to increase the 3-dB gain bandwidth [[19]]. The antenna achieves a 3-dB gain bandwidth of about 22.8%. However, the structure may increase the complexity of designing and fabricating. Typical PRSs for FPCAs are uniform periodic structures. These designs suffer from narrow bandwidth, while many of them still inflicted from the high profile. On the other hand, in the meta-lens researches, gradient metasurfaces are often used to achieve phase gradient [[20], [21]] or to construct a gradient refractive index [[22]-[24]], which improves the antenna gain. In the FPCA researches, there are rarely usages of gradient size metasurfaces. Based on the above discussion, there are few papers that can combine the advantages of low profile, wide −10 dB impedance bandwidth and wide 3-dB gain bandwidth in an FPCA. In order to overcome the above limitations, a new approach to enhance the bandwidth and the profile of FPCA is proposed by using X-direction gradient PRS and AMC in this paper. The operating mechanisms and theory of the approach are discussed and analysed. A wideband source antenna with a U-slot rectangular driven patch and three parasitic patches are proposed first. The proposed PRS is a novel complementary FSS structure, and a unit includes one hole and four small patches. The research has found that the field distributions of different frequencies are different on the PRS and the ground plane. Thus, the proposed PRS and the AMC have gradient sizes in X-direction, which meets a wide-band phase compensation. In addition, the gradient size AMC structure reduces the profile of the FPCA. 2 Theory and design 2.1 Theory analysis The schematic view of a conventional FPCA is shown in Fig. 1a. It is composed of a PEC ground plane and a conventional PRS placed, with a distance h. The electromagnetic waves experience multiple reflections and transmissions within the cavity. According to [[11]], the phase shift is introduced by the path length, the PEC ground plane and the PRS. The transmitted power can be derived by considering the interference of waves partially transmitted through the PRS. Hence, the directivity of the cavity at boresight is given as follows [[4], [10], [11]]: (1) where is the complex reflection coefficient of the PRS, PT is the transmitted power, PO is the power of the source, , and λ is the free-space wavelength. From (1), the magnitude of the reflection coefficient of the PRS is very important for gain stability versus frequency. The resonance condition can be easily derived by imposing the phase difference of the transmitted waves to be zero, , and it is written as follows [[5], [17]]: (2) or (3) where φPRS is the reflection phase of the PRS and φGND is the reflection phase of the PEC ground . Fig. 1Open in figure viewerPowerPoint Schematic view of the FPCA (a) Traditional FPCA, (b) Assumed FPCA field distribution When the reflection phase of the PRS is and N = 0, the cavity height of the antenna is about λ/2 at the operating frequency, which is a common design scheme in practical applications. However, according to formula (3), if the phase of PRS is not gradually increased with frequency, the resonance condition of FPCA can only be satisfied in very narrow frequency bands. Therefore, the traditional FPCAs have narrow 3-dB gain bandwidths though their gains are improved. In order to have phase increasing with frequency, multi-layer superstrates or metamaterials are typically employed on the FPCA [[2], [5], [17]]. However, these methods still have a narrow 3-dB gain bandwidth. The phase φPRS should be increased with frequency in a wide frequency band, which is difficult to achieve with traditional methods. In this research, it is assumed that the field distributions of PRS vary with frequencies, which means the characteristics of frequencies are dominated by different portions of the PRS as predicted in Fig. 1b. Therefore, a non-uniform PRS could be utilised to meet the wideband phase compensation condition, which was not studied before as far as we know. The method is based on the following phenomenon: the distribution of electrical energy density of conventional FPCA varies with frequencies, as shown in Fig. 2. As can be seen from the figure, with the increase of frequency, the electrical energy density on the PRS gradually spreads from the centre to the periphery. Therefore, it is proved that the method proposed in Fig. 1b is effective. According to the formula (3), different portions of the PRS require different phases φPRS to obtain wideband phase compensation condition. Thus, the PRS can be designed as a gradient structure. Fig. 2Open in figure viewerPowerPoint Electrical energy density of PRS at different frequency points in conventional FPCA (a) 6 GHz, (b) 7.5 GHz, (c) 9 GHz, (d) 10.5 GHz 2.2 Source antenna design To have a wideband FPCA, a wideband source antenna is required. In this research, microstrip antenna is chosen as the source element due to its low-profile, light-weight, ease of fabrication and feeding and directional radiation. However, the microstrip antennas are naturally narrow bandwidth [[25], [26]], typically 1–5%. In this design, a U-slot rectangular antenna is employed owing to its wide bandwidth [[27]]. In addition, the impedance bandwidth of the antenna is further improved by loading three closely parasitic patches around the U-slot rectangular driven patch to generate additional resonance [[28]] as shown in Fig. 3a. The substrate has a thickness of 4 mm and a relative dielectric constant of 2.2. Vertical slots are etched on the symmetrical parasitic patches to block lateral currents, which reduce the cross-polarisation and improve radiation characteristics. The antenna is fed through a 50 Ω coaxial cable. As shown in Fig. 3b, two more resonances are observed by the parasitic patches. In addition, the frequency of the U-slot rectangular patch is shifted to a higher frequency as the parasitic patches reduce the effective size of the U-slot patch. However, it is not a defect for compactness as the PRS would be large. Compared with the normal U-slot rectangular antenna, the 10-dB impedance bandwidth is improved from 11.4 to 58.1%. Its impedance bandwidth ranges from 6.1 to 11.1 GHz. In order to better compare the characteristics of the source antenna with the proposed FPCA, the gain and the radiation patterns of the source antenna are plotted in Figs. 13 and 18, respectively. Fig. 3Open in figure viewerPowerPoint Wideband source antenna (a) Top view of the source antenna (L1 = 17, L2 = 9, W = 9.2, W1 = 7, W2 = 8, W3 = 10, g = 0.5, all dimensions are in millimetres), (b) Reflection coefficients of the source antenna with and without the parasitic patches 2.3 Gradient PRS design As shown in Fig. 4, the proposed PRS has two metal layers of FSS, which are printed on the two sides of an F4-B substrate. The F4-B has a relative dielectric constant 2.6 and loss tangent 0.001. As shown in Fig. 4c, the top layer is a metal patch, whose size is gradually reduced on the x-axis, while the size of the patch is stable on the y-axis. The bottom layer is periodic square holes and the holes have an identical size as indicated in Fig. 4a. A detailed interpretation of using gradient PRS is explained later in the article. The schematic view of a unit cell of the PRS is shown in Fig. 5. As shown in the figure, a unit cell consists of four patches on the top and a square hole on the bottom. Fig. 4Open in figure viewerPowerPoint Gradient PRS structure applied to the design (a) Bottom side, (b) Side view, (c) Top side Fig. 5Open in figure viewerPowerPoint FSS unit cell (UC) with dimensions La = 9, LP = 10, p = 4.6–2.6 (gradually change in steps of 0.25, (all dimensions are in millimetres) The main goal of this gradient FSS is to achieve reasonable phase compensation in a wide frequency band. To characterise and investigate the scattering response of a unit, perfect electric and magnetic boundary conditions are applied along the y-directions and x-directions of the unit cell, respectively, and the structure is excited with a normal incident wave. The unit cell is simulated with CST Microwave Studio. The common complementary FSS structure, which only has one metal patch on the upper layer and a square hole at the bottom layer [[5]], is also investigated for comparison. The period of the common FSS is optimised as 8 mm. The size of the upper patch and the bottom square hole are 5.6 and 6.4 mm, respectively. As shown in Fig. 6a, compared with the common FSS, the proposed FSS has better transmission in a wide frequency band, while the S21 of the common FSS is small for the low-frequency band. Which means the proposed FSS would be a better choice for impedance matching in a wide frequency band of FPCA. The comparisons of the S11 curves of the antennas with the two FSS are given in Fig. 13a. It is found that the antenna with the proposed FSS has better impedance matching than the antenna with common FSS, especially at the low frequencies. Fig. 6Open in figure viewerPowerPoint Results of the FSS (a) Comparison of transmission coefficients between common FSS and Prop-FSS, (b) The proposed FSS reflection coefficients changes with the gradient P size Assuming the size of the FSS structure to be infinite, the directivity of an FPCA relative to that of the source antenna can be written as [[17]] (4) Where Dr is the relative directivity and ρ is the reflection magnitude of the FSS structure. It is found that a larger reflection ρ will lead to a higher directivity. However, as also indicated in Fig. 6b, the reflection ρ of an FSS structure is variable with frequency, and it can not be a constant reflection ρ in a wide frequency band. Thus, a traditional FPCAs can not obtain wide 3-dB gain bandwidths. As shown in Fig. 6b, the resonant frequency of the proposed FSS is increased with P decreasing, whereas the reflection amplitudes at the resonant frequencies of different P sizes are quite stable with the values around 0.45. Therefore, according to the formula (4), wide 3-dB gain bandwidth can be obtained by gradient FSS structure, and it can be estimated that the overall FPCA gain can be increased by 4–6 dBi based on the gain of the source antenna. The reflection phases of the proposed FSS are investigated as shown in Fig. 7. From the figure, by changing the length of P from 2.6 to 4.6 mm, the reflection phases are changing with frequency. It can be observed that the phases of different FSS sizes are gradually increasing with frequency at different frequency bands. Moreover, as the size P is changed from 4.6 to 2.6 mm, the rising phase-frequency band of the proposed FSS gradually moves from 6 to 12 GHz. This phenomenon is in good agreement with the theoretical schematic idea in Fig. 1b. That is, in order to meet the resonant conditions of the FPCA in the wider frequency band, the resonant conditions of different frequency bands should be satisfied in different positions of the PRS layer. Therefore, according to Formula (3) and Fig. 1b, the resonant conditions of the FPCA in the wider frequency band can be met by selecting the FSS structure with different P sizes. Fig. 7Open in figure viewerPowerPoint Proposed FSS reflection phase changes with the gradient P size at different frequency 2.4 Gradient AMC design An AMC is a periodic structure, which is basically an assembly of identical elements arranged in a one- or two-dimensional finite array. Various types of elements can be used as AMC, such as Mushroom-EBG, Jerusalem, UC-PBG and square patch [[29], [30]]. The square patch is used in our design as shown in Fig. 8. Due to its in-phase reflection characteristics, it is commonly used as a reflector or a ground to reduce the profile of the overall structure. Fig. 8Open in figure viewerPowerPoint AMC unit cell (UC) with dimensions a = 3.8–2.8 (gradually change in steps of 0.25), b = 4 (all dimensions are in millimetres) In the FPCA research, if the PEC ground in the cavity is replaced by AMC ground plane which introduces a φAMC phase shift, the profile of the antenna can be reduced. The (2) can be written as follows: (5) where φAMC and φPRS are the reflection phases of the PRS and AMC ground plane, respectively. According to the formula (5), as the φAMC is ∼0° and the φPRS is about , the profile of the FPCA can be reduced to λ0/4 [[11]], and the profile can be smaller than λ0/4 if the φPRS is smaller than . However, the narrow in-phase reflection bandwidth of a conventional AMC structure is difficult to meet the requirements of the proposed U-slot source antenna, who has an ultra-wideband (6–11.1 GHz). In general, the in-phase reflection band of the AMC is −90° to 90°. Moreover, according to the formula (5), h of the FPCA is calculated based on the actual phase of (φAMC + φPRS). Therefore, it is noted that a wideband FPCA requires a wide band of frequency increasing φPRS, and it would be better to have the φAMC also increase with frequency from (5). However, as indicated from a φAMC curve shown in Fig. 9, the φAMC is decreased with increasing frequency. Therefore, a uniform AMC surface is hard to achieve wideband FPCA. As also indicated in Fig. 9, by changing the length of the AMC patch from 3.8 to 2.8 mm, the φAMC = 0° point is gradually moved from 6.3 to 10.1 GHz. Therefore, different size patches can be used to maintain the broadband of φAMC = 0°, so that (φAMC + φPRS) increases with frequency. In this research, the size of the AMC structure is also gradually reduced in the x-axis direction and it is stable in the y-direction. Fig. 9Open in figure viewerPowerPoint Proposed AMC structure reflection phase changes with the gradient a size To further illustrate the causes of the usage of the X-gradient AMC structure, the electric energy density at different frequency points (6, 7.5, 9 and 10.5 GHz) of the X-gradient AMC structure are plotted in Fig. 10. The space in the middle is the location of the source antenna. It can be seen that, in the x-direction, the fields are spread out to the edges as frequency increases. It can be clearly seen that the energies of the 6 and 7.5 GHz are mainly concentrated on the centre of the AMC surface. While the energies of the 9 and 10.5 GHz gradually spread from the middle to the sides. However, in the y-direction, the fields remain stable for all frequencies. Therefore, the field distributions of the AMC surface indicate that the proposed AMC surface should be employed for the gradient size in the x-axis direction and stable dimensions in the y-direction. Fig. 10Open in figure viewerPowerPoint Electric energy density of the different frequency points of the proposed AMC ground (a) 6 GHz, (b) 7.5 GHz, (c) 9 GHz, (d) 10.5 GHz 3 Wideband FPCA designing According to the above discussions, the proposed FPCA is designed as shown in Fig. 11. The side view of the overall proposed FPCA is shown in Fig. 11a. Fig. 11b shows the pictures of the fabricated antenna. It consists of the source antenna, a PRS which is constructed by the complementary FSS and a square patch AMC surface. Both the PRS and AMC surfaces have gradient reduced size in the x-axis direction and stable dimensions in the y-direction. Four cylindrical hexagonal nylon spacers with relative permittivity of ɛr = 2.6 are embedded between the PRS and the source antenna to support the structures. These spacers are also taken into consideration in the simulations. The lateral dimension L of the antenna is 100 mmÍ100 mm (about 2.8λÍ2.8λ, where λ is the free space wavelength at the centre frequency 8.5 GHz). Ideally, according to the formula (5), when (φAMC + φPRS) is about 180°, the h should be about 8.2 mm (λ/4). However, the parameter h could be <8.2 mm owing to the following reasons, (φAMC + φPRS) is <180° in this case, the electromagnetic wave in the cavities are not plane waves, and the feeding antenna may also have an impact on the resonance frequency of the FPCA [[31]]. Fig. 11Open in figure viewerPowerPoint Geometry of the proposed antenna structure (a) Side view of the proposed antenna (h = 7.2, h1 = 1.6, h2 = 4, L = 100, all dimensions are in millimetres), (b) Manufactured prototype and measurement Therefore, the distance h between the ground plane and the PRS layer is studied by sweeping from 6.2 mm to 8.2 mm with a step of 1 mm as demonstrated in Fig. 12. Fig. 12a plots the S11 curves and Fig. 12b exhibits the gain curves. As shown in Fig. 12a, the impedance matching at a low frequency of the FPCA deteriorates at h = 8.2 mm, while the reflection coefficient curves exhibit very wide operating bands for all the h values. From Fig. 12b, the gain curves of the FPCA with different h values are higher than the source antenna. However, the gain of the FPCA with h = 8.2 mm is much lower than that of h = 6.2 and 7.2 mm. It is also shown that the antenna with h = 7.2 mm has optimal gain stability in the entire frequency band. Therefore, the height of the cavity is chosen as h = 7.2 mm. Fig. 12Open in figure viewerPowerPoint Investigations of the results in terms of the distance h (a) Reflection coefficient curves, (b) Gain curves The reflection coefficient and gain curves of the antennas are plotted in Fig. 13. Note that, the dielectric substrate FPCA utilised a dielectric substrate as PRS, with relative permittivity of 2.6 and thickness 1.6 mm. As shown in Fig. 13a, both the source antenna, the dielectric substrate FPCA and the proposed FSS FPCA have good impedance matching. However, as the common FSS structure is used, the matching in the 6 to 7 GHz band is deteriorated. Fig. 13Open in figure viewerPowerPoint Investigations of the antennas employing different structures of PRS (a) Reflection coefficient curves, (b) Gain curves As shown in Fig. 13b, the gain of the dielectric substrate FPCA is higher than the source antenna in the 6.5–8.2 GHz band. The increasing of the gain values is related to the thickness and dielectric constant of the dielectric substrate [[17]]. While the common FSS FPCA has the largest peak gain than the other two FPCAs. However, its 3-dB gain bandwidth is only 6.5 to 7.8 GHz (18.2%), which does not cover the entire impedance matching band. Compared with the previous two methods, the proposed FSS FPCA significantly improves the gain in almost the entire impedance matching frequency band. Its simulated 3-dB gain bandwidth ranges from 7.26 to 10.34 GHz (34.8%). Moreover, its gain is increased by about 4–7 dBi from 7 to 11.1 GHz. The antenna S11 and gain in terms of the uniform AMC (no gradient) and the proposed AMC (X-gradient) are investigated as demonstrated in Fig. 14. As shown in Fig. 14a, these two cases have good impedance matching. As shown in Fig. 14b, the proposed antenna with X-gradient AMC has better gain characteristics than the antenna with uniform AMC. The reason is that the uniform AMC units phase curve cannot be maintained the phase of (φAMC + φPRS) increasing with frequency in a wide band. Fig. 14Open in figure viewerPowerPoint Results of the FPCAs with uniform AMC and the proposed gradient AMC (a) Reflection coefficient variation curve, (b) Gain variation curve (the label X-gradient means the AMC is the gradient in x-direction and uniform along y-direction) The antenna S11 and gain in terms of the PRSs with Y-gradient, X/Y-gradient, no gradient and X-gradient (the proposed FSS) are investigated as demonstrated in Fig. 15. As shown in Fig. 15a. Good impedance matchings are obtained for all of the PRSs. As shown in Fig. 15b. The no gradient structure has a gain of 12 dBi around 7.5 GHz. However, its 3-dB gain bandwidth is narrow as the phase compensation band is narrow. While the gain enhancement of the Y-gradient structure is not high. The X/Y-gradient structure has better gain characteristics than the Y-gradient structure. Compared with the above three cases, the proposed X-gradient structure has best gain enhancement and stability. Fig. 15Open in figure viewerPowerPoint Performances of the FPCAs with different forms of PRS (a) Reflection coefficient curves, (b) Gain curves. The label Y-gradient means the FSS (PRS) is a gradient in y-direction and uniform along x-direction. The label X/Y-gradient means the FSS (PRS) is the gradient in both x- and y-directions. The label no gradient means the FSS (PRS) is uniform in both x- and y-directions (when the size of the Prop FSS upper four patches P = 3.6 mm) To further reveal the operating mechanisms of the proposed X-gradient FSS (PRS) and AMC, the electric energy densities at different frequency points (6, 8 and 10 GHz) of the proposed FPCA structure are plotted in Fig. 16. As shown in Fig. 16a, at the XOZ-plane, the electrical energy density of the propo
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